A novel reconfigurable microwave photonic (MWP) radar has been proposed and experimentally demonstrated. At a transmitting end, a microwave signal with a large bandwidth and ultra-low phase noise is generated by a Fourier domain mode locking optoelectronic oscillator. At a receiving end, photonics-based de-chirp processing is implemented by phase-modulating light waves in a dual-drive Mach-Zehnder modulator and mixing the modulated light waves at a photodetector. Without the requirement of external RF sources, the developed photonics-assisted programmable radar is capable of generating and processing microwave signals with adjustable format, bandwidth and central frequency. The proposed radar working from X to Ku band with an instantaneous bandwidth of 2 GHz is demonstrated. The reconfiguration of the radar is theoretically analyzed. The tunability of radar bandwidth and central frequency is investigated. Microwave imaging of a pair of trihedral corner reflectors based on the developed MWP radar is achieved.
© 2020 Optical Society of America under the terms of the OSA Open Access Publishing Agreement
A software-defined radar, where most of the processing, like signal generation, up- and down-conversion etc. is reconfigurable, is one of the most important developing directions of the next generation radars thanks to the capabilities of anti-jamming, flexibly setting the signal processing mode, and acquiring more detailed information of regions of interest [1,2]. A conventional fully electronically programmable radar usually consists of a series of microwave components operating efficiently only in a specific bandwidth or frequency band. Although a radar using direct digital synthesizers can achieve producing reconfigurable microwave waveforms, low output frequencies and large in-band spurious introduced by direct digital synthesizers restrain the scalability of the system function [3,4]. In contrast, microwave photonic (MWP) technologies provide promising solutions to break the bottleneck of microwave components and realize the reconfiguration of a radar, owing to the inherent advantages of photonics such as ultra-wide bandwidth, low transmission loss, wavelength agility and immunity to electromagnetic interferences [5–7]. Numerous researches on the reconfigurable microwave photonic radar have been studied intensively and impressive demonstrations have been achieved. Pro. Bogoni’s group in Italy presents the world’s first microwave photonic radar in 2014 [8–10]. By external-modulating a passive mode-lock laser (PMLL) and heterodyning two sidebands associated with two comb lines, the system can generate and process tunable signals without any electronic up- or down-conversion link. However, the limited pulse repetition rate of the PMLL, namely the comb-line spacing and the employment of optical/electrical bandpass filters to extract desired signals restrict the operation bandwidth and reconfiguration of the radar. To address this issue, a photonics-assisted central-frequency-programmable and bandwidth-tailorable radar architecture is proposed [11–13]. The generation of microwave signals is based on spectral filtering and heterodyning pre-chirped pluses of a PMLL via wavelength-to-time mapping. Both the central frequency and bandwidth of the generated waveforms can be independently controlled by changing the parameters of optical filters. The received echoes are optically stretched in the time domain and compressed in the frequency domain, avoiding the use of electronic down-conversion links or high-speed analog-to-digital converters (ADCs). Thanks to the large bandwidth of stretched pulses, the operation bandwidth can be as large as tens of gigahertz. Nevertheless, due to the limited dispersion during wavelength-to-time mapping, the pulse duration is only tens of nanoseconds, which leads to a low time-bandwidth product (TBWP). To acquire radar signals with a large TBWP, a microwave photonic radar employing photonic frequency-multiplying and photonic stretch processing is developed and demonstrated, broadening signal’s bandwidth and reducing the sampling rate of the ADC [14–19]. Operation bandwidth, central frequency and pulse duration of the proposed MWP radar can be changed by programming a low-frequency, narrow-bandwidth RF source. However, the de-chirp processing structure based on devices integrated with several Mach-Zehnder modulators (MZMs), suffers from bias voltage control complexity and bandwidth limitation induced by electrical 90° RF hybrids. Although a structure based on a phase modulator followed by an optical tunable filter can also implement de-chirp processing, the optical filter limits the bandwidth scalability of the radar . In addition, the RF source has large in-band distortions and a low output frequency, affecting the quality of generated signals and the reconfigurable capabilities of radars. To circumvent the problems introduced by RF sources, a photonics-based radar utilizing a logic-operation-based photonic digital-to-analog converter (PDAC) is reported [20–22]. An up-converting single tone achieved by an optical frequency comb and a broadband intermediate frequency signal produced by the PDAC are mixed to generate wideband signals with tunable central frequencies and a broad bandwidth with no aid of electrical wideband microwave sources. However, the radar signal has a low signal-to-noise ratio associated with the low bit number of the PDAC. In addition, the high-speed code pattern generators in the PDAC increase the operation complexity. Due to the dependence on electrical RF sources or intrinsic limitations of signal generation mechanisms, the aforementioned tunable MWP radars have insufficient reconfiguration and are suitable for limited applications.
In this letter, a new reconfigurable photonics-based radar that is capable of transmitting and receiving microwave signals without any RF sources, is proposed and experimentally demonstrated. In the transmitter, Fourier domain mode locking (FDML) technique is employed in a conventional tunable optoelectronic oscillator (OEO) to generate a wideband transmission signal. In the receiver, a light wave is separately phase-modulated by an incoming echo signal and a reference signal in a dual-drive Mach-Zehnder modulator (DDMZM), the sidebands are then mixed in a photodetector (PD) to realize de-chirp processing. The significant feature of the developed MWP radar is no requirement of RF sources, resulting in high reconfigurability in regards to signals’ central frequency, bandwidth and format. Experimentally, an MWP radar operating from X to Ku band with a bandwidth of 2 GHz and a period of 5 µs is presented. The reconfigurable capability of the radar is proved by theoretical analysis. Performances associated with signals’ frequency agility, phase noise, compression ratio, and bandwidth tailorable are investigated. Besides, an experiment of imaging a pair of trihedral corner reflectors (TCRs) is conducted, verifying the effectiveness and expected precision of the system.
2. Principle of operation
The schematic diagram of the proposed MWP radar is shown in Fig. 1. The transmitter of the radar consists of a frequency-sweeping laser, a phase modulator (PM), an optical notch filter (ONF), an optical delay line, a PD1, two low noise amplifiers (LNA1 and LNA2), two electrical wideband bandpass filters (EBPF1 and EBPF2), a three-port power divider (Div), a phase shifter and a power amplifier (PA1).
In a conventional OEO, there is usually a time-invariant filter. When a filtered signal propagates along the loop and returns the filter, and a multiple of 2π phase variation is added, the signal will oscillate if the loop gain is larger than unity . Due to the filter being time-invariant, only a single tone can exist in the loop. While in an FDML-OEO scheme, a time-variant filter is employed. A specific frequency component in the loop passes through the filter only if the filter at the exact time is tuned at the same spectral position. Thus, the phase variation condition for oscillation is changed with time, leading to amounts of longitude modes oscillate in turn during the loop round-trip delay in the cavity. Therefore, a wideband signal can be achieved and its response of frequency varying with respect to time can be controlled by changing the spectrum-time relation of the filter [24–27]. In the developed FDML-OEO-based transmitter end, the frequency-sweeping laser, the PM, the ONF and PD1 in the OEO loop form a transfer function equivalent to a time-variant microwave photonic filter (MPF) [25,28–30]. The oscillation frequency can be controlled by changing the frequency detuning between the central frequency of the laser and the central frequency of the ONF. By adjusting the driving current waveform of the frequency-sweeping laser to a saw-tooth function, triangle function and exponential function, the time-variant MPF is changed correspondingly and a wideband signal with frequency varying linearly, triangularly and exponentially with time can be achieved, respectively.
In the receiver end, the de-chirp structure is composed of a fiber laser, an DDMZM, a PD2, an ADC, a digital signal processor (DSP), two electrical bandpass filters (EBPF3 and EBPF4), an LNA3 and a PA2. A light wave from the fiber laser is fed into the DDMZM, which is integrated with two phase modulators. In the DDMZM, the incident light wave is split into two arms. The light waves in both arms are phase-modulated by a reference signal and an incoming echo signal, respectively. At the output of PD2, a de-chirped signal is generated. Subsequently, the signal is sampled by the ADC and Fourier transformed by the DSP to acquire the targets’ information. As can be seen, the de-chirp structure avoids using RF hybrids so that the instantaneous bandwidth of the signal that the de-chirp structure can process is only determined by the bandwidth of DDMZM and PD2.
Theoretical analysis of the photonics-based de-chirp receiver is presented as follows. For the sake of simplicity, assuming that the reference signal is the same as the transmission signal and expressed as
The de-chirp processing can be achieved when the DDMZM is working at the maximum or minimum transmission point. Without loss of generality, only one situation that the DDMZM works at the minimum transmission point is considered. When the light waves in both arms are combined and detected by PD2, the output microwave signal can be expressed as
Range resolution is an index that measures the ability of a radar system to distinguish two targets at range. When the de-chirped signal is Fourier transformed in the range direction, the theoretical range resolution of the radar is 10), the larger bandwidth of the signal, the higher range resolution can be obtained.
To sum up, the developed MWP radar is capable of working flexibly without any RF sources, allowing to process programmable microwave signals with large frequency ranges and broad bandwidths.
3. Experiments and results
Firstly, a radar transmitter based on an FDML-OEO is developed. A light wave centered at 1550.09 nm with a power of 9 dBm from a homemade frequency-sweeping laser is sent to the PM, whose 3-dB bandwidth is 20 GHz and half-wave voltage is around 5 V. The modulated light wave is then delayed by a dispersion-free optical fiber (residual dispersion: −0.189 ps/nm) of around 1 km. A phase-shifted fiber Bragg grating with a tuning resolution of 250 MHz is applied as the ONF, whose output is detected by PD1 with responsivity of 0.7 A/W and 3-dB bandwidth of 35 GHz. After detection, a microwave signal is generated and amplified by around 50 dB using electrical amplifiers. EBPF1 is employed to suppress undesired harmonic components induced by the nonlinearity of the amplifiers. The Div splits the generated signal into three parts. One part is phase-shifted and fed to the PM to form a closed loop. Another part is used as a reference signal for de-chirp processing in the radar receiver. The third part is amplified by PA1, filtered by EBPF2 and transmitted by a transmitting antenna as a transmission signal.
The frequency responses of the MPF in the FDML-OEO are recorded by a vector network analyzer. When the central frequency of the ONF is adjusted from 1550.16 to 1550.24 nm, frequency responses of the MPF in X band and Ku band are shown in Figs. 2(a) and 2(b), respectively. Figure 2(c) shows a zoom-in view of the measured frequency response with a central frequency of 14.6 GHz. It is observed that the 3-dB bandwidth of the MPF is about 110MHz. Also, the stability of the FDML-OEO has been investigated. The system is capable of stably operating in a room temperature for about 10 minutes with no significant power fluctuations. As to the frequency stability, due to that the oscillation frequency is sensitive to the wavelength drift of the laser and spectrum drift of the ONF, a frequency shift of a few kHz can be observed. Given that the time duration of radar detection is usually less than 10 seconds, the measured frequency shift has negligible effect on radar detections. If a temperature controller is applied in the system, the stability will be further improved.
Subsequently, a wideband microwave signal is generated by the developed transmitter and frequency agility of the transmitter is investigated. The measured longitudinal mode frequency interval of the OEO loop is about 203 kHz. To generate an LFMCW signal, the driving signal of the homemade frequency-sweeping laser is designed as a saw-tooth waveform with a repetition frequency of about 203 kHz. When the oscillator is stable, a broadband LFMCW signal is produced. By tuning the central frequency of the ONF, the central frequency of the wideband signal can be changed. The spectra of signals with the same bandwidth of 2 GHz but different central frequencies are shown in Fig. 3, demonstrating that the proposed RF-source-free radar has the capability of generating signals in both X and Ku bands. The power variations are mainly introduced by the unflatten frequency responses of the MPF.
In addition, characteristics associated with phase noise, pulse compression ratio (PCR) and linearity of the wideband signal are evaluated. Phase noise is an important metric that degrades the ability to process Doppler information in radar systems and degrades error vector magnitude in digital communication systems. It can be described in many ways, the most common is single side band phase noise and defined as the ratio as the power density at an offset frequency from the carrier to the total power of the carrier signal. In an OEO scheme, the phase noise is usually characterized by the Yao-Maleki equation 11) indicates that the oscillator’s phase noise is independent of the oscillation frequency and so that the OEOs can be applied to generate high-frequency signals with low phase noise. In the experiment, the measured phase noise is −118 dBc/Hz at a 10-kHz offset frequency when the oscillation frequency is 15 GHz, as shown in Fig. 4. The peaks that have a frequency spacing corresponding to a free spectral range of the OEO result from the nondominant oscillation modes and can be suppressed by utilizing a multi-loop OEO structure [32,33]. Notice that each frequency component of the wideband signal is generated through oscillation in the same OEO loop. Thus, the phase noise of the measured single tone can be treated as a typical value of the wideband signal. For comparison, phase noise performance of a commercial RF source (Tektronix AWG70001A) is also investigated. The phase noise of the signal generated by the OEO at a 10-kHz offset frequency is 20 dB lower than that generated by the AWG. The phase noise close to the carrier is relatively high because the OEO is sensitive to ambient environmental changes.
Figure 5(a) shows the generated microwave waveform in the time domain, which is measured by a high-speed digital oscilloscope. The period of the microwave signal is about 5 µs, which is equal to the OEO loop time delay. Figure 5(b) shows the measured spectrum of the generated microwave waveform with a span of 600 kHz. It can be seen that the measured frequency detuning between two adjacent modes is about 203 kHz, consistent with the repetition rate of the driving current signal. The PCR is defined as the ratio of signal pulse width before and after compression. It determines the improvement of signal-to-noise ratio of after compression. As for a LFMCW signal, the PCR is approximately equal to the TBWP. Figure 5(c) reports the result of cross-correlation between the 1st and the 21st pulses. As can be seen that the 3-dB pulse width of the compressed pulse is 0.44 ns, corresponding to a PCR of about 11189. Due to inherent limitation in the generation of an FDML-OEO scheme, the longest pulse duration of the microwave signal is equal to the loop time delay . Therefore, the PCR and TBWP of the signals are not as good as those of programmable digital radar transmitters whose pulse duration is theoretically unlimited. However, if a longer fiber is used in the system, the PCR, TBWP and phase noise can be further improved. The peak-to-sidelobe ratio is around 12.6 dB, and the 0.3-dB power discrepancy between first sidelobes of the compressed pulse is mainly caused by the nonlinearities in the LFMCW signal. While the linearity of an LFMCW signal is one of the most important metrics influencing the quality of radar images [34,35]. It is defined as
To investigate the linearity of the produced LFMCW waveform, the calculated instantaneous frequency-time diagram is shown in Fig. 5(d). As can be observed that obvious nonlinearities exist in the signal, which is caused by the residual phase. The residual phase is necessarily obtained for nonlinearity compensation, and can be acquired by mixing an ideal reference LFMCW signal with the tested signal in the digital domain. Figure 5(e) shows the residual phase in the time domain. To further evaluate the linearity of the signal, a plot illustrating signal’s frequency deviation varying with time is depicted in Fig. 5(f). The largest frequency deviation of the signal is around 156 MHz, corresponding to a linearity of 7.8%.
To observe the repeatability of the signals, microwave waveforms in different period (1st, 50th, 100th) in the time domain are shown and superposed in Fig. 6(a). A zoom-in view of the dashed frame shown in Fig. 6(a) with a 60-ns time duration is plotted in Fig. 6(b). It can be seen that small initial phase jitters which are mainly caused by the imperfect wavelength repetition of the laser exist in the produced signal. As for our radar system with de-chirp architecture, the small initial phase jitters have negligible impact on the radar detection. If a laser with higher wavelength stability is applied, the initial phase stability can be further enhanced.
Next, a programmable MWP radar is built up based on the setup shown in Fig. 1. Bandwidth tailorable of the radar and process capability of the de-chirp structure are verified by a point target detection test. A fiber optical delay line module (Emcore OTS-ODLS) is employed to simulate a point target located at 5.5 m far away from the radar. As shown in Fig. 7(a), by varying the slope of the driving signal and tuning the central frequency of the ONF, a microwave transmission signal centered at 17.5 GHz with a bandwidth of 1 GHz is generated. The transmission signal is then sent to the fiber optical delay line module and directly applied to the receiver for de-chirp processing. In the receiver, the echo signal is filtered, amplified and launched into one RF port of the DDMZM (Fujutsu FTM7937EZ) with a bandwidth of 35 GHz, while the reference signal is sent to the other RF port. The modulated optical carriers at the output of the DDMZM are applied into PD2 whose 3-dB bandwidth is 18 GHz and responsivity is 0.65 A/W and recorded by the ADC. Figure 7(b) shows the electrical spectrum of the de-chirped echo after Fourier transform in the range direction. As we can see, some false targets appear due to the nonlinearities of the transmission signal. Therefore, a nonlinearity compensation method is necessary. Given that different targets correspond to different time delays and nonlinearity compensation algorithm in the time domain only works for a target with a fixed distance, the residual phase varying with frequency is used for nonlinearity compensation so that the impact of different time delays can be circumvented. The nonlinearity compensation is carried out as follows: firstly, convert the recorded de-chirped signal and transmission signal to complex signals through Hilbert transform. Secondly, a complex echo signal is recovered by mixing the conjugation of converted de-chirped signal and the transmission signal in the time domain. Thirdly, by mixing the converted transmission signal and the conjugation of an ideal complex reference LFMCW signal in the frequency domain, the residual phase varying with respect to frequency is acquired. Subsequently, in the frequency domain, the phase of the echo signal is compensated by using the residual phase to improve the linearity of the echo signal. Finally, by mixing the conjugation of the phase-compensated echo signal and the ideal reference signal in the time domain, a linearity-compensated de-chirped signal is acquired and after Fourier transform, excellent pulse compression along the range direction is achieved which is depicted in Fig. 7(c). A distinct peak at about 5.4 m is found, the peak-to-sidelobe ratio equals to 13.3 dB, and the measured range resolution is around 14.8 cm, which is consistent with the theoretical value.
Finally, a proof-of-concept rotational microwave imaging experiment where the proposed MWP radar operates in Ku band is conducted. As shown in Fig. 8, (a) rotating platform is located at around 6.7 m far from the radar, a pair of trihedral corner reflectors (TCRs) is placed on the rotating platform with a speed of 26.5 degree/s for rotational imaging. Two TCRs are put 0.35 m away in the range direction, and 1.15 m away in the cross-range direction. A wideband LFMCW transmission signal with a bandwidth of 2 GHz (13.4-15.4 GHz) is emitted by the FDML-OEO-based transmitter, reflected by the TCRs and de-chirped in the photonics-based receiver.
As expected, without linearity compensation, the electrical spectrum of the de-chirped echo of the rotating TCRs shown in Fig. 9(a) contains not only real targets but also false targets. After linearity compensation, false targets are suppressed and two distinct targets can be found in Fig. 9(b). The measured range resolution is around 7.2 cm. An ISAR image of the two targets is obtained after two-dimensional Fourier transforming the linearity-compensated de-chirped signal, as depicted in Fig. 9(c). The calculated distances of the two TCRs are 34.8 cm in the range direction and 98.6 cm in the cross-range direction, which are consistent with the real condition.
Notice that there are mainly two reasons for the nonlinearity of the signal: 1) the intrinsic frequency modulation response nonlinearity of the frequency-sweeping laser vs. the driving current makes the periodical spectrum variation of the MPF not ideal; 2) the poor temperature stability of the OEO make it easy for mode hopping during the oscillation. If a feedback optoelectronic loop and a temperature control device are employed, the nonlinearity of the signal can be further mitigated so that a linearity compensation algorithm is no longer needed [36–38].
The experimental results show the effectiveness of the developed MWP radar. Compared with existing radar structures in proposed in [8,11,16,21], the FDML-OEO-based radar has a potential to achieve excellent reconfigurability and a large bandwidth thanks to the RF-source-free setup. Therefore, the proposed MWP radar scheme is an attractive alternative for developing multifunctional and reconfigurable high-resolution radar systems.
In summary, we have presented, theoretically analyzed and experimentally demonstrated a novel RF-source-free reconfigurable MWP radar. An MWP radar operating from X to Ku band with a bandwidth of 2 GHz and a period of 5 µs is presented. Characteristics of the generated signal are evaluated and the reconfigurability of the radar is confirmed. Besides, an experiment of imaging a pair of trihedral corner reflectors (TCRs) is conducted, demonstrating that the MWP radar has great potential in reconfigurable radar applications.
National Key Research and Development Program of China (2018YFA0701900, 2018YFA0701901); National Natural Science Foundation of China (61690191, 61701476).
The authors thank Qizhuang Cen and Tengfei Hao for their assistance with experiments.
The authors declare no conflicts of interest.
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