Record-high 19.25Gb/s real-time end-to-end dual-band optical OFDM (OOFDM) colorless transmissions across the entire C-band are experimentally demonstrated, for the first time, in reflective electro-absorption modulator (REAM)-based 25km standard SMF systems using intensity modulation and direct detection. Adaptively modulated baseband (0-2GHz) and passband (6.125 ± 2GHz) OFDM RF sub-bands, supporting signal line rates of 9.75Gb/s and 9.5Gb/s respectively, are independently generated and detected with FPGA-based DSP clocked at only 100MHz as well as DACs/ADCs operating at sampling speeds as low as 4GS/s. The two OFDM sub-bands are electrically multiplexed for intensity modulation of a single optical carrier by an 8GHz REAM. The REAM colorlessness is experimentally characterized, based on which optimum REAM operating conditions are identified. To maximize and balance the signal transmission performance of each sub-band, on-line adaptive transceiver optimization functions and live performance monitoring are fully exploited to optimize key OOFDM transceiver and system parameters. For different wavelengths within the C-band, corresponding minimum received optical powers at the FEC limit vary in a range of <0.5dB and bit error rate performances for both baseband and passband signals are almost identical. Furthermore, detailed investigations are also undertaken of the maximum aggregated signal line rate sensitivity to electrical sub-band power variation. It is shown that the aforementioned system has approximately 3dB tolerance to RF sub-band power variation.
© 2013 OSA
Wavelength-division-multiplexed passive optical networks (WDM-PONs) have widely been considered as one of the most promising strategies for satisfying the exponentially increasing end-users’ demands for broadband services, and an excellent solution for future mass deployment in next generation PONs (NG-PONs) such as NG-PON2 and beyond, since WDM-PONs offer significantly improved signal transmission capacity, enhanced system flexibility and wavelength scalability, virtual point-to-point connection enabling each individual end-user to be granted a single wavelength, protocol transparency and excellent network security [1,2]. To practically deploy WDM-PONs, the main challenges are cost-effectiveness and colorlessness of optical network units (ONUs). To address such challenges, several techniques have been proposed, which include, for example, spectrum-slicing of super luminescent diodes (SLDs) or optical amplifiers [3,4], wavelength tunable lasers in ONUs [5,6] and remote modulation of reflective intensity modulators [7,8].
Among all the aforementioned techniques proposed previously, reflective intensity modulator-based ONUs are very attractive for WDM-PONs since the implementation of centrally-controlled dynamic wavelength management in an optical line terminal (OLT) allows the reflected upstream signal transmission from the ONU in a wavelength agnostic manner. More importantly, since ONU-side laser sources are not present, the ONU transceiver complexity is greatly simplified and consequently the overall WDM-PON installation cost is significantly reduced by a factor proportional to the total number ONUs accommodated simultaneously in the network. To achieve reflective intensity modulation-based cost-effective ONUs, reflective intensity modulators that can be utilized include, for example, reflective semiconductor optical amplifiers (RSOAs) [9–11], reflective electro-absorption modulators (REAMs) [12–14] and reflective Fabry-Perot lasers [15–17]. Among these reflective intensity modulator candidates, REAMs can be easily scaled for data line rates higher than 10Gbits/s compared to other reflective modulators due to their high modulation bandwidths. For example, 40Gb/s transmission over 20km standard single mode fiber (SSMF) with offline digital signal processing (DSP) has been achieved by using a 20GHz SOA-REAM.
To further enhance the performance and flexibility of WDM-PONs and still maintain their compatibility with existing time-division-multiplexed (TDM) PONs, optical orthogonal frequency division multiplexing (OOFDM) has been considered as one of the strongest contenders for practical applications , since OOFDM has inherent and unique advantages including adaptive provision of hybrid dynamic bandwidth allocation (DBA) in both the frequency and time domains, significant reduction in network complexity, adaptability to imperfect transmission system characteristics and potential for cost-effective mass deployment. It is, therefore, preferable if use is made of OOFDM signals in REAM-based ONUs in WDM-PONs incorporating SSMF systems utilizing simple intensity modulation and direct detection (IMDD).
To exploit the feasibility of WDM-PONs incorporating colorless REAM-based ONUs, we have experimentally achieved 10Gb/s real-time end-to-end transmissions of single-band OOFDM (SB-OOFDM) signals over 25km SSMF . However, to further improve the transmission capacity of a colorless REAM-based SB-OOFDM ONU in a cost-effective approach, system designers face great challenges, as higher speed digital-to-analogue converters (DACs) and analogue-to-digital converters (ADCs) are required, this makes it difficult to avoid a prohibitive ONU transceiver cost. Another important issue to consider when employing ultra-high sampling speed DACs/ADCs in the SB-OOFDM transceivers is the escalation of the data bus capacity between the converters and the DSP logic circuits. These ultra-high capacity data buses can be challenging to implement when trying to meet tight cost and power constraints.
By employing a multi-band OOFDM (MB-OOFDM) approach, where multiple OFDM signals are frequency division multiplexed (FDM) in the electrical domain, virtually all the aforementioned disadvantages associated with the SB-OOFDM approach are eliminated. Firstly, the bandwidth requirements of key components such as DACs/ADCs are dictated only by the sub-band bandwidth and are no longer dependent on the total system capacity. The increased flexibility in component choice can be highly beneficial for greatly lowering the ONU transceiver cost. Secondly, by careful selection of the sub-band transmission capacity, the ONU performance can be designed to meet the required peak user transmission capacity whilst maintaining sufficient intra-band DBA functionality. In addition, MB-OOFDM also has the salient advantage of significantly reducing transceiver complexity in terms of DSP logic requirements. The reduction in DSP complexity is particularly pertinent to the fast Fourier transform (FFT) and inverse FFT (IFFT) algorithms, which benefit significantly from the reduced number of subcarriers processed in the case of MB-OOFDM.
Furthermore, the reduced DSP complexity combined with the reduced sampling speeds and logic clock speeds can also result in a significant reduction in ONU power consumption. In addition, by designing MB-OOFDM optical transceivers that are capable of multiplexing/demultiplexing any OFDM sub-band also brings a unique advantage of improving network operation flexibility and introducing dynamic traffic management at sub-wavelength granularity without requiring extra expensive optical devices such as reconfigurable optical add/drop multiplexers (ROADMs). Recently, by making use of low-speed 4GS/s DACs/ADCs, we have reported a real-time end-to-end dual-band OOFDM transmission system operating at 19.125Gb/s over 25km SSMF in an electro-absorption modulated laser (EML)-based IMDD system .
In this paper, record-high 19.25Gb/s end-to-end real-time REAM-based colorless dual-band OOFDM transmissions across the C-band are experimentally demonstrated, for the first time, in simple 25km SSMF IMDD transmission systems. Optimum REAM operating conditions are identified for supporting the colorless operation of the dual-band OOFDM signals over the C-band. Moreover, the system performance robustness is also extensively explored to determine the system tolerance to sub-optimum sub-band RF power. It is shown that aggregate system capacity is maintained almost constant for sub-band power variations as large as 3dB due to the utilization of adaptive bit and power loading.
2. Real-time dual-band OOFDM REAM-based colorless transmission system setup
The real-time end-to-end dual-band OOFDM REAM-based transmission system setup is shown in Fig. 1 with all key component/system parameters detailed in Table 1. Independent digital and RF electronics are employed in the two separate transmitters for the simultaneous generation of two separate OFDM sub-bands. One sub-band with an adaptively loaded signal bit rate of 9.75Gb/s occupies the spectral region from 0 to 2GHz and is referred to as a baseband OFDM signal. To generate the second sub-band with an adaptively loaded signal bit rate of 9.5Gb/s, a second 0-2GHz OFDM baseband signal amplitude modulates a 6.125GHz RF carrier, generating a double sideband OFDM signal occupying the spectral region from 4.125 to 8.125GHz. This is referred to as a passband OFDM signal. A single receiver is employed to receive either OFDM sub-band with the inclusion of an RF down-conversion stage for the passband.
For both OFDM sub-band transceivers, the employed field programmable gate array (FPGA)-based real-time DSP is identical to that implemented in [20,21]. The transmitter DSP design consists of the following major functions: pseudo-random binary test data generation, pilot-tone insertion, on-line adaptive subcarrier modulation using bit and power loading of 15 data-carrying subcarriers with modulation formats taken from 16-quaternatry amplitude modulation (QAM), 32-QAM or 64-QAM, 32-point IFFT for the generation of real-valued OFDM time-domain symbols, on-line adaptive clipping and 8-bit sample quantization, addition of 25% cyclic prefix and parallel-to-serial conversion for sample output to a 4GS/s, 8-bit DAC.
The functions implemented in the corresponding real-time DSP in the receiver [20,21] are outlined below: serial-to-parallel conversion of OFDM signal samples from a 4GS/s 8-bit ADC, detection of pilot-subcarriers, automatic symbol alignment and cyclic prefix removal, 32-point FFT for the generation of complex frequency domain subcarriers from the received real-valued time domain symbols, channel equalization, on-line adaptive demodulation of the 15 data-carrying subcarriers and bit error rate (BER) analysis of total sub-band BER and individual subcarrier BERs. On-line performance monitoring of the receiver measured BERs, system frequency responses and subcarrier constellations is achieved through the FPGA’s embedded logic analyzer function. This allows instant analysis of the system’s transmission performance which, combined with the on-line control of transmitter DSP parameters, RF signal gains, REAM operating conditions and optical launch power, provides rapid optimization of the system performance.
As shown in Fig. 1, at the transmitter side, a 9.75Gb/s real-valued baseband OFDM signal emerging from the corresponding DAC is first amplified and its power is adjusted appropriately via variable electrical attenuators (VEAs). Simultaneously, a 9.5Gb/s passband RF OFDM signal is also generated by first amplifying an OFDM signal from another DAC, up-converting with a 6.125GHz RF carrier via a double-balance mixer to produce a double-sideband RF signal, then passband-filtering to attenuate unwanted out-of-band signals and passing through a second RF gain stage with variable signal power achieved by VEAs. The relative OFDM RF sub-band power levels and the absolute RF dual-band power level can therefore both be controlled with a resolution of 1dB to allow optimum power levels to be determined. The two sub-bands with a total signal bit rate of 19.25Gb/s are finally combined in a 6dB resistive RF coupler. To reduce the effect of imperfect RF impedance matching at the coupler, RF isolators are placed at the corresponding coupler inputs .
After passing through an optical circulator with a 1.4dB insertion loss, an optical continuous wave (CW) supplied by a tunable laser source (TLS) is injected, at an optical power of 3dBm, into a polarization-insensitive REAM modulator with an electrical modulation bandwidth of ~8GHz. The REAM modulator is a buried heterostructure MQW device with an anti-reflection coating on the front facet and a high-reflection coating on the rear facet. Detailed REAM descriptions and parameters can be found in . The power-optimized dual-band 1.48Vpp electrical analog OFDM signal and a wavelength-dependent optimum reverse bias DC voltage are combined in a 50GHz bias-T to modulate the CW optical wave of a selected wavelength in the REAM modulator operating at a TEC controlled temperature of 20°C. An erbium doped fiber amplifier (EDFA) at the REAM output allows control of optical launch power which is optimized to 8.8dBm for launching into the 5dB-loss 25km SSMF.
At the receiver side, the optical signal passes through a variable optical attenuator (VOA) to adjust the received optical power (ROP) and is then directly detected by a 12.4GHz PIN photodetector with integrated transimpedance amplifier (TIA) to convert the received dual-band OOFDM signal into the electrical domain.
To receive the baseband OFDM signal, the RF down-conversion section shown in Fig. 1 is omitted, whilst to receive the passband OFDM signal, the RF down-conversion section is included. In the RF down-conversion circuit, a bandpass filter is utilized to extract the passband OFDM signal from the dual-band OFDM signal before down-conversion to baseband. The received baseband signal or down-converted passband signal passes to a variable RF gain section consisting of a fixed RF gain amplifier and VEAs. This is followed by an anti-aliasing low pass filter (LPF) and balun to generate the differential signal required by the 4GS/s 8-bit ADC. The receiver’s RF gain is manually adjusted according to the ROP to maintain an optimum peak-to-peak signal level at the ADC input. In practice this can be achieved by an automatic gain control (AGC) circuit. The digital signal emerging from the ADC is directed to the receiver’s FPGA to proceed with the inverse DSP processes compared to the transmitter side, and also to perform automatic symbol alignment, channel estimation and equalization functions. For both mixers in the passband transmitter and receiver sides, the local oscillator (LO) signals are derived from the same signal source. A variable delay line is employed to correctly align the phase of the receiver’s LO with the phase of the received carrier. In practical implementation, an automatic OOFDM synchronization technique previously developed in  may be modified to enable the compensation of phase mismatches between the transmitter and the receiver. In addition, the relative and absolute OFDM sub-band power levels are also fully optimized independently in combination with on-line optimization of the REAM operating conditions, adaptive loading profiles and clipping levels of each sub-band. The optimized transmitter side parameters and operating conditions are fixed when system performance measurements are made for each sub-band.
Making use of the OOFDM transceiver design and the experimental system setup illustrated in Fig. 1, optimizations of the transceiver and system operating conditions are first conducted in real-time to maximize the obtainable signal line rate by effectively compensating for the system frequency response roll-off effect through adaptive bit and power loading performed on the information-bearing subcarriers [19–21, 24,25]. Figure 2(a) illustrates the obtained optimum subcarrier bit allocation profiles corresponding to a total raw signal bit rate of 19.25Gb/s, of which 15.4Gb/s can be employed to carry user data because of the use of a 25% cyclic prefix. It can be seen in Fig. 2(a) that lower signal modulation formats tend to occur in the lower and higher frequency subcarriers, this is attributed to the combined effects of strong subcarrier intermixing upon direct detection in the receiver and the residual system frequency response roll-off. The direction detection-associated subcarrier intermixing effect is pronounced for subcarriers having low frequencies, and the residual system frequency response roll-off effect is pronounced for subcarriers having high frequencies. In addition, the occurrence of relatively lower signal modulation formats in some central subcarriers, as shown in Fig. 2(a), is mainly due to the imperfect subcarrier orthogonality-induced subcarrier intermixing effect. Furthermore, to identify the wavelength-dependent reverse bias DC voltage required by the REAM intensity modulator, optical power loss characteristics of the REAM modulator against reverse bias DC voltage are also measured for representative wavelengths within the C-band, the corresponding experimental results are depicted in Fig. 2(b). Throughout the present paper, Fig. 2(b) is used as an important reference for selecting optimum REAM modulator operating conditions.
3. Experimental results
3.1 System BER performance at 1550nm
The system frequency responses of the REAM-based 19.25Gb/s real-time dual-band OOFDM 25km SSMF IMDD system at 1550nm for each sub-band are presented in Fig. 3(a). The measurements are made from the transmitter IFFT input to the receiver FFT output and normalized to the first subcarrier power. For the passband transmission, an effective system frequency response is determined as the effect of passband transmission-induced relative subcarrier attenuation is included. This indicates that Fig. 3(a) does not reveal the true system frequency response of the RF and optical channels in the passband frequency region of 4.125-8.125GHz. From Fig. 3(a), the maximum system frequency response roll-offs of approximately 22dB and 23dB for the baseband and passband, respectively, are observed, which are mainly attributed to the characteristic sin(x)/x response of the zero-order hold DAC, the on-chip filtering of the DAC outputs and the contributions of the RF electronics .
The measured sub-band BER performances against ROP are shown in Fig. 3(b) for system configurations of both the optical back-to-back and the entire 25km SSMF system at a wavelength of 1550nm. It can be seen in Fig. 3(b) that the 9.75Gb/s baseband and the 9.5Gb/s passband have similar BER developing trends and their ROP differences at the adopted forward error correction (FEC) limit of 2.3 × 10−3 are < 0.3dB. These performance similarities confirm the effectiveness of utilizing adaptive bit and subcarrier/sub-band power loading in combating the high system frequency response roll-off effect, as shown in Fig. 3(a). In addition, to achieve similar BER performance for each sub-band, it is also necessary to employ a passband to baseband RF power ratio of 1.91dB at the transmitter. After transmission through the 25km SSMF, the passband to baseband RF power ratio is reduced to −1.08dB, indicating that the passband OOFDM signal suffers more attenuation than the baseband OOFDM signal. This is due to the effects of limited REAM modulation bandwidth and IMDD-induced channel fading. Both these effects cause an increase in system frequency response roll-off with increasing subcarrier frequency. Figure 3(b) also shows that the passband performs slightly better than the baseband at the high ROP region, this is because, in comparison with the passband signals, the inter-sub-band interference effect has a stronger impact on the baseband signals, as observed and explained later in the subsection.
Based on the adaptively loaded bit profiles in Fig. 2(a) and subcarrier power profiles in Fig. 3(a) as well as the other aforementioned optimum settings, Fig. 4 shows the corresponding subcarrier error distributions for both the baseband and passband OFDM sub-bands after 25km SSMF transmission measured at their maximum ROPs in Fig. 3(b) and a wavelength of 1550nm. In Fig. 4 it is clearly observed that the use of adaptive bit and power loading can result in almost uniform BER distributions across the entire subcarrier range, whose BER variations are as low as ± 3.9% for the baseband and ± 5.2% for the passband.
To explore the impact of inter-sub-band interference on minimum ROP after 25km SSMF transmission at 1550 nm, the BER performance of each sub-band is measured with the other sub-band being switched off in the DSP but with all corresponding components still connected and powered. Experimental measurements in Fig. 5 show that the inter-sub-band interference induced power penalty to the passband OOFDM transmission at the adopted FEC limits is less than 0.5dB, whilst the corresponding power penalty to the baseband OOFDM transmission is 1.3dB. This indicates that, for the present system, the inter-sub-band interference has relatively strong impact on the baseband OOFDM signal. Such observed power penalties are in close agreement with those reported in . The physical reason underpinning the above-mentioned behaviors are that unwanted intermixing frequency products generated upon square-law photon detection in the receiver are predominantly located in the baseband spectral region .
3.2 Colorlessness performance across the C-band
To experimentally examine the colorlessness of the considered system across the C-band, in addition to the adoption of the adaptive bit loading profiles presented in Fig. 2(a), for each sub-band adaptive power loading is also performed on-line for all the information-bearing subcarriers as shown in Fig. 6 for three representative wavelengths within the C-band. Figure 6 depicts the loaded subcarrier power profiles, their corresponding received subcarrier power profiles, as well as the system frequency responses, all normalized to the first subcarrier for baseband and passband. For the three representative wavelengths, the optimized reverse DC bias voltages are 1.75V for 1560nm, 1.45V for 1550nm and 0.85V for 1536nm according to Fig. 2(b). The electrical driving voltage is almost wavelength independent thus a single optimized driving voltage of 1.48Vpp is taken throughout the paper. As expected, Fig. 6 shows that, for all these wavelengths considered, the maximum system frequency response roll-offs of ~23dB are measured for both sub-bands, which agree very well with those mentioned in Section 3.1. This indicates that the system frequency response is independent of the wavelengths considered.
Making use of the optimized REAM operating conditions, and the optimum bit and power loading profiles shown in Fig. 2(a) and Fig. 6, Fig. 7 shows the measured total channel BER performances for the different wavelengths of the OOFDM signal transmission over 25km SSMF for 9.75Gb/s baseband and 9.5Gb/s passband sub-bands. As seen in Fig. 7, for all of these wavelengths across the entire C-band, colorless transmission is achievable with wavelength-dependent ROP variations of <0.5dB at the FEC limit for both sub-bands. Such slight residual wavelength-dependent minimum ROP variations originate mainly from an optical wavelength-induced variation in extinction ratio of the REAM modulated signal .
Received constellations of representative subcarriers recorded prior to channel equalization for these wavelengths are plotted in Fig. 8(a) for the baseband and Fig. 8(b) for the passband. The constellations are measured at their minimum sub-band BERs after transmission over 25km SSMF. For all these constellations, large variations in subcarrier amplitude levels are clearly seen, as expected from the measurements presented in Fig. 6. In addition, these constellations show very little deviations for different wavelengths, this is verified by the small BER differences between these wavelengths, as shown in Fig. 7.
3.3 System robustness to RF sub-band power variations
From discussions in Sections 3.1 and 3.2, it is clear that adaptive bit and subcarrier/sub-band power loading provides a simple and effective means of compensating for the system frequency response roll-off effect. This provides the dual-band OOFDM transmission system with considerably enhanced flexibility to achieve the best overall system performance, which is always maximized according to the actual component/system characteristics. As a direct result, this offers the dual-band OOFDM system a capability of combating both individual component and system variations. For example, for a REAM-based dual-band OOFDM system with an optimized combination of electrical sub-band powers, the technique enables system designers to maximize the aggregated signal line rate. On the other hand, for a system with a specific minimum aggregated signal line rate, the technique allows a tolerance to the electrical sub-band power deviation from its optimum value. This section is dedicated to addressing this issue, for the first time.
The following experimental procedures are adopted in examining the aforementioned issue: Firstly the electrical sub-band power is set to the specific optimized value as determined in Section 2, then the passband (baseband) RF power is decreased in 1dB increments whilst maintaining the power of the baseband (passband) constant. The maximum signal line rate is then obtained for each new sub-band power level by making full use of the on-line adaptive bit and power loading, corresponding to which the total channel BER of each sub-band is kept smaller than the FEC limit for each of the sub-band power combinations. For all measurements in this section, optical wavelength is fixed at 1550nm. It should be noted that sub-band differential power levels cannot be increased above the optimum level as this leads to the REAM being driven in the non-linear operating region.
In Fig. 9(a), the passband RF power is fixed at its optimum value and the baseband RF power is decreased, the baseband signal line rate decreases gradually with decreasing baseband RF power due to a reduction in the corresponding modulation index. The baseband signal rate drops form 9.75Gb/s to 8.5Gb/s for a 6dB RF power decrease, while the passband signal line rate is slightly increased form 9.5Gb/s to 9.75Gb/s mainly because of the partial alleviation of the inter-sub-band interference. This leads to the total dual-band signal line rate dropping from 19.25Gb/s to 18.25Gb/s when there is a 6dB baseband power differential below the optimum value.
A similar trend is also observed for an opposite case as shown in Fig. 9(b) where the passband signal line rate drops form 9.5Gb/s to 8.25Gb/s for a 5dB RF power differential, while the baseband signal line rate is increased form 9.75Gb/s to 9.875Gb/s. When the sub-band power differential reaches 6dB, the passband BER cannot be lower than the FEC limit even with 16-QAM modulation format taken on all subcarriers (7.5Gb/s), as indicated by the blue dash line in Fig. 9(b). Therefore, the total dual-band signal line rate drops from 19.25Gb/s to 17.375Gb/s with a 5dB passband power differential below the optimum value. Comparing Figs. 9(a) and 9(b), the passband signal line rate drops more quickly than the baseband signal line rate for an identical sub-band power differential, indicating that the passband signal suffers more distortions than the baseband signal during transmission.
It is also very interesting to note that the total dual-band OOFDM signal line rate can be maintained at >19Gb/s and kept reasonably balanced between two sub-bands for sub-band power differentials as large as 3dB. Thus the system exhibits at least a 3dB tolerance to RF sub-band power differential.
REAM intensity modulators have been utilized, for the first time, to experimentally demonstrate colorless, dual-band, real-time, end-to-end, IMDD OOFDM transmission incorporating 25km SSMF. Adaptively modulated 9.75Gb/s baseband (0-2GHz) and 9.5Gb/s passband (6.125 ± 2GHz) OFDM RF sub-bands are electrically multiplexed for intensity modulation of a single optical carrier by the REAM. The colorlessness of the REAM has been characterized, based on which optimum REAM operating conditions are identified. In the aforementioned system architecture, 19.25Gb/s colorless dual-band transmission of end-to-end real-time OOFDM signals have been successfully achieved for various representative wavelengths across the entire C-band. Over such a wavelength window, minimum ROPs at the FEC limit vary in a range of less than 0.5dB. Making use of on-line sub-band power loading, detailed investigations of electrical sub-band power differential-dependent aggregated signal line rates have also been undertaken, which show that the aforementioned system has at least 3dB tolerance to RF sub-band power differential. This work thus confirms the potential of employing REAM-based dual-band OOFDM ONUs for NG-PONs.
It should be pointed out that, to further improve the aggregated OOFDM signal transmission capacity, system flexibility and transceiver cost-effectiveness, the dual-band transceiver has been extended to a 4GS/s DAC/ADC-based tri-band architecture, which results in experimental demonstrations of 25.25Gb/s real-time OOFDM transceivers in our research lab. Our experimental measurements also indicate that a further increase in the number of OFDM sub-bands is also feasible. The aforementioned new experimental results will be reported elsewhere in due course.
This work was supported by the PIANO + under the European Commission’s ERA-NET Plus Scheme within the project OCEAN under Grant Agreement 620029. The work of Zhang and Wang was supported in part by the National Natural Science Foundation of China (61132004).
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