We experimentally demonstrate heterodyne coherent detection of 8 × 112-Gb/s ultra-density wavelength-division-multiplexing (WDM) polarization-division-multiplexing quadrature-phase-shift-keying (PDM-QPSK) signal after 1120-km single-mode fiber-28 (SMF-28) transmission. The spectral efficiency (SE) is 4b/s/Hz. It is the first time to realize WDM signal transmission with high SE by adopting heterodyne coherent detection. At the heterodyne coherent receiver, intermediate frequency (IF) down conversion is realized in digital frequency domain after analog-to-digital conversion. A digital post filter and 1-bit maximum likelihood sequence estimation (MLSE) adopted after carrier phase estimation (CPE) in the conventional digital-signal-processing (DSP) process is used to suppress the enhanced noise and crosstalk as well as overcome the filtering effects. The bit-error ratio (BER) for all channels is under the forward-error-correction (FEC) limit of 3.8 × 10−3 after 1120-km SMF-28 transmission.
© 2013 OSA
With the development of large-bandwidth and high-speed electronic analog-to-digital converters (ADCs) and photo detectors (PDs), very recently, coherent detection with digital signal processing (DSP) has been attracting a great deal of interest in research community once again [1–8]. It’s well known that coherent detection includes homodyne detection and heterodyne detection. Homodyne detection has been intensively discussed and investigated, and already applied in commercial or lab-demonstration coherent communication systems for 100G, 400G, 1T or beyond [3–5] with high spectral efficiency (SE). Particularly, by adopting the technique of post filter and 1-bit maximum likelihood sequence estimation (MLSE), polarization-division-multiplexing quadrature-phase-shift-keying (PDM-QPSK) signal transmission, with the single-channel bit rate up to 100G, 200G and 400G, has been demonstrated in the homodyne coherent system with SE of 4b/s/Hz [6–8]. In this paper, we will demonstrate that the same SE can be obtained in the heterodyne coherent system if we adopt the same technique of post filter and 1-bit MLSE.
The previous reports for heterodyne detection include 310-Mb/s QPSK heterodyne system using a fourth-power optical phase-locked loop , limited 5-Gb/s 4-ary quadrature amplitude modulation (4-QAM) coherent optical transmission over 20km  and limited 20-Mbaud 64- and 128-QAM coherent optical transmission over 525km . Compared to homodyne detection, by down-converting in-phase (I) and quadrature (Q) components to the intermediate frequency (IF) at the same time, not only can heterodyne detection reduce the number of balanced PDs (BPDs) and ADCs of coherent receiver into half, but also there is no need to consider the delays between I and Q components in the PDM signal. The conventional dual-hybrid structure also becomes unnecessary. Besides, we have experimentally demonstrated that, for the wavelength-division-multiplexing (WDM) system adopting heterodyne detection, one local oscillator (LO) laser can be used for two neighboring WDM channels, and thus the number of LO lasers can be reduced into half compared to homodyne detection . As a result, heterodyne detection is much more hardware-efficient than homodyne detection. Furthermore, based on large-bandwidth PDs and ADCs, IF down conversion, I/Q separation and equalization can be realized by DSP in digital domain after analog-to-digital conversion [13, 14]. Thus, heterodyne detection has more advantages over homodyne detection in terms of integration and parallel processing, and can be widely applied to various optical transport and access networks. However, the required bandwidth of ADC used for heterodyne detection is twice of that used for homodyne detection. Furthermore, compared to homodyne detection, there is an extra 3-dB signal-to-noise ratio (SNR) penalty for heterodyne detection. That’s why all commercially available products for coherent transmission systems at present adopt homodyne detection instead of heterodyne detection.
In this paper, we propose and experimentally demonstrate the generation, transmission and heterodyne detection of 8 × 112-Gb/s WDM PDM-QPSK signal over 1120-km single-mode fiber-28 (SMF-28) with SE of 4b/s/Hz. It is the first time to realize WDM signal transmission with high SE by adopting heterodyne coherent detection. Two BPDs, two ADCs, two polarization beam splitters (PBSs) and two optical couplers (OCs) are needed in this heterodyne coherent detection. IF down conversion is realized in digital frequency domain after analog-to-digital conversion. A digital post filter and 1-bit MLSE adopted after carrier phase estimation (CPE) in the conventional DSP process is used to suppress the enhanced noise and crosstalk as well as overcome the filtering effects. The bit-error ratio (BER) for all channels is under the forward-error-correction (FEC) limit of 3.8 × 10−3  after 1120-km SMF-28 transmission. On the other hand, the BTB BER for the WDM channel without post filter and 1-bit MLSE is over 1 × 10−2, which means that we cannot realize error-free transmission even with advanced FEC.
2. Principle of digital post filtering and heterodyne coherent detection
The idea of duo-binary signaling or correlative coding , which has only 1-bit memory length and is a specific class of partial response signaling, is to introduce a controlled amount of inter-symbol interference (ISI) into the signal instead of trying to eliminate it completely. The introduced ISI can be compensated in digital domain at the receiver. Thus, the ideal symbol-rate packing of 1 baud per Hertz per polarization can be achieved without the requirements for unrealizable filters based on the Nyquist theorem. Thus, it is necessary for multi-symbol optimal decision schemes, such as maximum-a-posteriori-probability (MAP) and MLSE, to take advantage of symbol correlation existing in the received partial response signals. The challenge is that the number of states and transitions grows exponentially with the involved memory length. For instance, the MLSE length of 10 means 410 states and 411 transitions in lane-dependent PDM-QPSK signals  and thus computational complexity significantly increases in practical implementation. Meanwhile, in the bandwidth-limiting optical coherent system, the noise in high frequency components of signal spectrum and the crosstalk are both enhanced after linear equalization algorithm, such as classic constant modulus algorithm (CMA) .
A linear digital delay-and-add finite-impulse-response (FIR) filter with a transfer function of H(z) = 1 + Z−1, when added after CPE in the conventional DSP process at the coherent receiver, can provide a simple way to achieve partial response, which will effectively mitigate the enhanced crosstalk and noise . At the same time, the MLSE algorithm, with a significantly reduced memory length, is employed to realize symbol decoding and optimal decision. From the constellation point of view, the effect of the post filter transforms 4-point QPSK signal into 9-point quadrature duo-binary one, as illustrated in Fig. 1 . As a result of the delay-and-add effect, the 2-ary amplitude shift keying (2-ASK) I and Q components disappear and are then independently converted into two 3-ASK symbol series. The mechanism for the generation of ‘9-QAM’ signals can be considered as the superposition of the two 3-ASK vectors on a complex plane [19–21]. The size of constellation points represents the relative number of points generated after the post filter.
The principle of heterodyne coherent receiver and IF down conversion in digital frequency domain can be found in . The heterodyne coherent receiver includes two PBSs for polarization-diversity splitting between the received optical PDM signal and the LO, two OCs, two BPDs and two ADCs.
3. Experimental setup and results for heterodyne coherent WDM transmission
Figure 2 shows the experimental setup for the generation and transmission of 8 × 112-Gb/s WDM PDM-QPSK signal on a 25-GHz grid with heterodyne coherent detection as well as post filter and 1-bit MLSE.
At the transmitter, eight external cavity lasers (ECLs), with linewidth less than 100kHz and maximum output power of 14.5dBm, are divided into two groups and respectively used as the continuous-wavelength (CW) light source for the odd and even channels. Each group of ECLs has 50-GHz neighboring frequency spacing. The odd and even groups of ECLs are independently combined by two polarization-maintaining OCs (PM-OCs), and then modulated by two I/Q modulators (I/Q MODs). Each I/Q MOD is driven by a 28-Gbaud electrical binary signal, which, with a pseudo-random binary sequence (PRBS) length of 215-1, is generated from an electrical dual-channel pulse pattern generator (PPG). For optical QPSK modulation, the two parallel Mach-Zehnder modulators (MZMs) in each I/Q MOD are both biased at the null point and driven at the full swing to achieve zero-chirp 0- and π-phase modulation . The phase difference between the upper and lower branches of each I/Q MOD is controlled at π/2. The subsequent polarization multiplexing for each path is realized by polarization multiplexer, comprising a PM-OC to halve the signal into two branches, an optical delay line (DL) to provide a 150-symbol delay, an optical attenuator to balance the power of two branches and a polarization beam combiner (PBC) to recombine the signal. The programmable wavelength selective switch (WSS) on a 25-GHz grid is used to combine and spectrally shape the odd and even channels. Insets (a) and (b) respectively show the optical spectra before and after the WSS for the odd channel, while insets (c) and (d) the even channel. The WDM signal is launched into the straight line of 14 spans of 80-km SMF-28. Each span has 18-dB average loss and 17-ps/km/nm chromatic dispersion (CD) at 1550nm without optical dispersion compensation. Erbium-doped fiber amplifier (EDFA) is used to compensate for the loss of each span. The total launched power (after EDFA) into each span is 8dBm, corresponding to ~-1dBm per channel at 112Gb/s. Insets (e) and (f) show the optical spectra for all channels before and after 1120-km SMF-28 transmission, respectively.
At the receiver, a tunable optical filter (TOF) with 3-dB bandwidth of 0.2nm is used to choose the desired channel. An ECL with linewidth less than 100kHz is used as the LO, which has 18.5-GHz frequency offset relative to the fifth channel at 1549.34nm. Two PBSs and two OCs are used to implement polarization diversity of the received optical signal together with the LO in optical domain before balanced detection . Inset (g) shows the X-polarization optical spectrum after polarization-diversity splitting. It can be seen that the frequency spacing and power difference between the LO and the received optical signal is 18.5GHz and 20dB, respectively. A band-pass electrical amplifier (EA), with 31-dB gain and 15~32-GHz frequency range, is used after each BPD. The analog-to-digital conversion is realized in the real-time digital storage oscilloscope (OSC) with 80-GSa/s sampling rate and 30-GHz electrical bandwidth. Inset (h) shows the electrical spectrum for the X-polarization component centered on 18.5GHz obtained after analog-to-digital conversion. For the detailed DSP after analog-to-digital conversion, firstly, the received signals are down-converted to the baseband by multiplying synchronous cosine and sine functions, which are generated from a digital LO . Secondly, a T/2-spaced time-domain FIR filter is used for CD compensation, where the filter coefficients are calculated from the known fiber CD transfer function using the frequency-domain truncation method. Thirdly, two complex-valued, 13-tap, T/2-spaced adaptive FIR filters, based on classic CMA, are used to retrieve the modulus of the PDM-QPSK signal and realize polarization de-multiplexing. The subsequent step is carrier recovery, which includes residual frequency-offset estimation and CPE. The former is based on fast Fourier transform (FFT) method while the latter fourth-power Viterbi-Viterbi algorithm. A post filter is then adopted to convert the binary signal to the duo-binary one. The final symbol decision is based on 1-bit MSLE [19–21]. Finally, differential decoding is used to eliminate π/2 phase ambiguity before BER counting. In this experiment, the BER is counted over 10 × 106 bits (10 data sets, and each set contains 106 bits).
In the following part, ch5 and CH5 are respectively used to denote the 28-Gbaud PDM-QPSK single channel and WDM channel both at 1549.34nm. All experimental measurements are carried out in the case of 18.5-GHz frequency offset. Figure 3 shows BTB BER versus optical SNR (OSNR) for ch5 and CH5, respectively. Post filter and 1-bit MLSE is adopted for CH5. Compared to ch5, there is about 1.3-dB OSNR penalty at the BER of 3.8 × 10−3 for CH5. On the other hand, the BTB BER for the WDM channel without post filter and 1-bit MLSE is over 1 × 10−2, which means that we cannot realize error-free transmission even with advanced FEC. Inset (a) shows the Y-polarization BTB error-free constellation for CH5 after CPE, while inset (b) after further post filtering. Figure 4(a) gives BTB BER versus 3-dB bandwidth of TOF for CH5 when the OSNR is set at 24dB. The optimum BER performance is attained when the 3-dB bandwidth of TOF is 17.9GHz. Insets (I) and (II) show the X-polarization BTB constellations for CH5 after CPE and further post filtering, respectively. The corresponding 3-dB bandwidth of TOF and BER is 17.9GHz and 7 × 10−4, respectively. Figure 4(b) gives BER/OSNR versus transmission distance for CH5. There is 10-dB OSNR decrease when the transmission distance increases from 400km to 1120km. Meanwhile, the corresponding BER increases from 1.5 × 10−4 to 2.5 × 10−3 due to reduced OSNR. Figure 5(a) gives BER/OSNR versus total input power into fiber for CH5 after 1120-km SMF-28 transmission. The optimum BER performance is attained when the total input power is 9dBm (corresponding to ~-1 dBm per channel), and the corresponding OSNR is 20.8dB. The BER increases when the total input power is over 9dBm. It’s because, as the total input power increases, the enhanced fiber nonlinearity becomes dominant and worsens the BER performance. Figure 5(b) shows the BER of all channels after 1120-km SMF-28 transmission. The BER for each channel with the optimum input power of −1dBm/channel is under 3.8 × 10−3. Insets (I) and (II) respectively show the X- and Y-polarization constellations for CH5 over 1120-km transmission after CPE, while insets (III) and (IV) after further post filtering. The corresponding BER is 3 × 10−3.
We experimentally demonstrate the generation and transmission of 8 × 112-Gb/s WDM PDM-QPSK signal on a 25-GHz grid with heterodyne detection as well as post filter and 1-bit MLSE. The BER for all channels is under the FEC limit of 3.8 × 10−3 after 1120-km SMF-28 transmission. On the other hand, the BTB BER for the WDM channel without post filter and 1-bit MLSE is over 1 × 10−2, which means that we cannot realize error-free transmission even with advanced FEC. Our experimental results show that the technique of post filter and 1-bit MLSE can improve SE and receiver sensitivity and is also useful in the heterodyne coherent system.
This work was partially supported by NNSF of China (61107064, 61177071, 60837004, 61250018), NHTRDP (863 Program) of China (2011AA010302, 2012AA011302), NKTR&DP of China (2012BAH18B00) and ICPSSTA of Shanghai (12510705600).
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