We report a polarization-multiplexed, 10 Gsymbol/s 64 QAM coherent transmission over 150 km using an optical voltage controlled oscillator (OVCO). The OVCO enables us to realize a low phase noise optical phase-locked loop (OPLL) due to its wideband operation independent of the frequency modulation (FM) bandwidth of an LD. As a result, 120 Gbit/s, 64 QAM data were successfully transmitted over 150 km with a power penalty as low as 1 dB.
© 2013 Optical Society of America
A spectrally efficient digital coherent transmission technology with a multi-level modulation is making rapid progress with the aim of expanding the transmission capacity within a limited optical amplification bandwidth . Of many modulation formats, multi-level quadrature amplitude modulation (QAM) is advantageous as regards expanding the spectral efficiency toward the Shannon limit. QAM has been utilized for a number of coherent optical transmission experiments, and 1024 QAM transmission has recently been achieved with a spectral efficiency of 13.8 bit/s/Hz . To realize coherent transmission with a high multiplicity, carrier-phase synchronization between transmitted data and a local oscillator (LO) is indispensable. A conventional scheme involves carrier-phase estimation based on digital signal processing where LO feedback control is not needed [3, 4]. However, as the multiplicity and the symbol rate increase, a high-speed and large-scale electronic circuit is required, resulting in greater computational complexity. Moreover, the accuracy of carrier-phase estimation becomes insufficient for higher-order QAM.
On the other hand, an optical phase-locked loop can provide carrier-phase synchronization with high accuracy and potentially with low power consumption. OPLLs have already been demonstrated by using fiber lasers, and successfully used to demonstrate 1024 QAM signals. Laser diodes (LDs) with a narrow linewidth of several tens of kHz have also been employed for OPLL to demodulate BPSK  and QPSK  signals. We previously reported an LD-based OPLL circuit that employs a frequency-stabilized external cavity laser diode (ECLD) with a linewidth of 4 kHz, and demonstrated a 120 Gbit/s, polarization-multiplexed (Pol-Mux), 10 Gsymbol/s, 64 QAM coherent transmission over 150 km . However, in these experiments, it was difficult to achieve the low phase noise OPLL operation required for a higher-order QAM. This is because the feedback loop bandwidth was limited by the frequency modulation (FM) bandwidth of the LO, where the phase of the LO was controlled by changing the injection current or external cavity length with a piezoelectric transducer (PZT). In our previous experiment, an error floor was observed in the bit error rate (BER) characteristics under a back-to-back condition even with a high optical signal-to-noise ratio (OSNR).
To overcome this bottleneck, a sub-carrier OPLL system with an optical voltage controlled oscillator (OVCO) has been proposed [8, 9]. The OVCO make a wideband OPLL operation possible without limiting the FM response bandwidth of the LO. Recently, we applied this scheme to our previous OPLL circuit and achieved low phase noise OPLL operation .
In this paper, we utilize our new OPLL system with an OVCO, and successfully demonstrate a 120 Gbit/s, Pol-Mux, 10 Gsymbol/s, 64 QAM coherent transmission over 150 km. By using the OVCO, the BER characteristics were greatly improved compared with those obtained with a previous OPLL system .
2. Experimental setup for 120 Gbit/s, 64 QAM coherent optical transmission employing OVCO
Figure 1 shows our experimental setup for a 120 Gbit/s, polarization-multiplexed (Pol-Mux), 64 QAM coherent transmission with an OVCO. The coherent CW laser we used was a 1538.8 nm C2H2 frequency-stabilized ECLD with a linewidth of 4 kHz . The output of the laser was split into two arms. In one arm, it was modulated by an IQ modulator, which was driven with a 10 Gsymbol/s, 64 QAM signal generated by an arbitrary waveform generator (AWG). Here, a pre-equalization process was adopted to compensate for the distortions, which were caused by individual components such as the AWG and the IQ modulator. Furthermore, the nonlinear phase rotation caused by self-phase modulation (SPM) during transmission was also pre-compensated. We introduced SPM compensation using software at the AWG by adding a phase shift,Eq. (2), N is the number of spans and PI and PQ are the optical powers of the I and Q data. Pol-Mux was performed with a polarization beam combiner (PBC).
In parallel, the output of the transmitter was coupled to an optical frequency-shifter (OFS) consisting of an intensity modulator and an optical filter with a bandwidth of 300 MHz, where its frequency was downshifted by 10 GHz (fc - 10 GHz) against the carrier frequency. This signal was used as a pilot tone signal for the OPLL in the receiver. Here, the polarization of the pilot tone was the same as that of the Y-polarization data signal. Figure 2 shows the optical spectra of the QAM data and pilot tone signals measured with a 0.01 nm resolution bandwidth. These signals were transmitted over a 150 km dispersion-managed fiber link. Each span of the fiber link consisted of a 50 km super large area (SLA) fiber with a dispersion of 19.5 ps/nm/km and a 25 km inverse dispersion fiber (IDF) with a −40 ps/nm/km dispersion. The average loss of the transmission fiber was 17 dB.
Our receiver is composed of two parts. One is a polarization-diversity coherent receiver, and the other is an OPLL circuit. In the coherent receiver, the transmitted QAM data signal was amplified and then homodyne detected with an OVCO signal whose phase was locked to the data signal via the pilot tone by using a polarization-diverse 90-degree optical hybrid and four balanced photo-detectors (B-PDs). Finally, the detected data signals were A/D-converted using a high-speed digital oscilloscope (40 Gsample/s, 16 GHz bandwidth) and demodulated with software in an offline condition. We calculated the BER from 123 kbit demodulated signals.
The OPLL circuit consists of an OVCO, a photo-diode (PD), an RF amplifier, a double balanced mixer (DBM), a synthesizer, and a loop filter with a bandwidth of approximately 10 MHz. The OVCO is composed of an ECLD with a linewidth of 4 kHz as an LO, a LiNbO3 (LN) optical phase modulator, a tunable optical filter with a bandwidth of 6 GHz, and a 10 GHz RF-VCO. In the OVCO, the output signal of the ECLD (fLO) is phase modulated by the LN phase modulator driven by the RF-VCO (fmod = 10 GHz) with a modulation depth of 0.46π, and then the first longer-wavelength sideband of the modulated signal is extracted with the tunable optical filter. The frequency and phase of the OVCO signal (fOVCO = fLO-fmod) was changed by applying a voltage to the RF-VCO. In the OPLL circuit, the phase of the beat signal between the extracted pilot tone and the OVCO signals (IF signal: fIF = |fpilot-fOVCO|) was compared with that of a reference signal generated from the synthesizer (fsyn) at 10 GHz by the DBM. The voltage phase error signal from the DBM was fed back to the RF-VCO through a lag-lead loop filter.
In Fig. 3(a) the gray and red curves show the optical spectra of the OVCO before and after filtering, respectively. After filtering, extra sidebands are sufficiently eliminated with a side-mode suppression ratio of more than 35 dB. Figure 3(b) shows the delayed self-heterodyne spectrum of the OVCO with a measurement resolution of 2 kHz . The linewidth was approximately 4 kHz, which is the same as that of the LO. Figure 3(c) shows the frequency tuning characteristics of the OVCO. The frequency was tuned with a tuning ratio of 1.67 MHz/V. Figure 3(d) shows the FM response and phase characteristics of the OVCO. The 3 dB bandwidth was approximately 4 MHz, which was the same as that of the RF-VCO.
3. Experimental results
Figure 4 shows the BER of the demodulated 10 Gsymbol/s, 64 QAM signal after a 150 km transmission for various powers launched into each fiber span. From these results, the launchpower was optimized to −1 dBm, where the optical power of the QAM data and the pilot tone signal were −4 dBm/pol and −14 dBm, respectively. The optical spectra of the data signal before and after 150 km transmission are shown in Fig. 5. The OSNR was degraded from 43 to 27 dB during the 150 km transmission.
Figures 6(a) and 6(b) show the beat spectrum between the OVCO and the pilot tone as an intermediate frequency (IF) signal under OPLL operation with 2 and 20 MHz spans, respectively. The linewidth of the spectrum was less than 10 Hz, which was below the measurement resolution. The phase noise was widely suppressed, which resulted in a low noise IF signal with an SNR of approximately 60 dB. Figure 7 shows the single sideband (SSB) phase noise spectrum. This graph also shows the phase noise spectrum of the reference signal generated from a synthesizer. The phase noise variance (RMS) of the IF signal estimated by integrating the SSB noise power spectrum from 10 Hz to 1 MHz was only 0.6 degrees. In comparison with the performance of our previous OPLL where the LO phase was tuned by changing the injection current , the phase noise was greatly reduced from 5 to 0.6 degrees. This is due to the wideband OPLL operation with the OVCO. On the other hand, when we compare this result with the reference signal, the phase noise of the IF signal is 3 times larger than that of the reference signal (0.2 deg.). In particular, the phase noise in the high frequency region (100 kHz~1 MHz) is large. We can expect to suppress the phase noise of the IF signal further by increasing the OVCO bandwidth using a wideband RF-VCO.
Figures 8(a) and 8(b) show constellation maps for the 10 Gsymbol/s, 64 QAM signal under back-to-back and after a 150 km transmission at OSNRs of 43 and 27 dB, respectively. After the transmission, the constellation points were broadened due to the OSNR degradation.
Figure 9 shows the BER characteristics as a function of the OSNR at a 0.1 nm resolution bandwidth back-to-back and after a 150 km transmission. This graph also shows the BER characteristics obtained with the previous OPLL circuit with gray curves . In comparison to the previous results, there is no error floor in the BER characteristics under the back-to-back condition with the OVCO. Furthermore, the power penalty at a BER of 2x10−3 after transmission was successfully reduced from 3 to 1 dB.
We successfully transmitted a polarization-multiplexed, 10 Gsymbol/s, 64 QAM (120 Gbit/s) signal over 150 km with a low power penalty of 1 dB. This result was achieved by the use of OVCO with a low phase noise OPLL operation. The present coherent transmission scheme is expected to be a candidate for future multi-level coherent transmission systems, especially those with higher-order multiplicity.
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