Photonic signal processing offers the advantages of large time-bandwidth capabilities to overcome inherent electronic limitations. In-fibre signal processors are inherently compatible with fibre optic microwave systems that can integrate with wireless antennas, and can provide connectivity with in-built signal conditioning and electromagnetic interference immunity. Recent methods in wideband and adaptive signal processing, which address the challenge of realising programmable microwave photonic phase shifters and true-time delay elements for phased array beamforming; ultra-wideband Hilbert transformers; single passband, widely tunable, and switchable microwave photonic filters; and ultra-wideband microwave photonic mixers, are described. In addition, a new microwave photonic mixer structure is presented, which is based on using the inherent frequency selectivity of the stimulated Brillouin scattering loss spectrum to suppress the carrier of a dual-phase modulated optical signal. Results for the new microwave photonic mixer demonstrate an extremely wide bandwidth operation of 0.2 to 20 GHz and a large conversion efficiency improvement compared to the conventional microwave photonic mixer.
© 2013 OSA
Photonic signal processing offers a new, powerful paradigm for processing high bandwidth signals. It opens up new possibilities for overcoming the inherent bottlenecks caused by limited sampling speeds in conventional electrical signal processors . The motivation arises from the potential of exploiting the unique, high time-bandwidth product capabilities of photonic signal processing [1–3]. In-fibre signal processors are inherently compatible with fibre optic microwave systems, and can provide connectivity with in-built signal conditioning. These new techniques transcend the limitations of existing electronic methods, and enable new structures to be realised, which not only can process high-speed signals but which can also realise adaptive operation.
Microwave photonic signal processing promises to bring new capabilities to radio frequency (RF) and millimeter-wave systems. A key benefit of using optical techniques for wideband electromagnetic systems is that the entire RF/mm-wave spectrum constitutes only a small fraction of the carrier optical frequency, thus offering the fundamental advantage of very little frequency-dependent loss and dispersion across entire microwave bands of interest. Photonic signal processing leverages the advantages of the optical domain, which can then benefit from the wide bandwidth, low loss, and natural immunity to electromagnetic interference (EMI) that photonics offers. Photonic signal processing also enables dynamic reconfiguration of the processor characteristics over multi-GHz bandwidths, which is a difficult task for electrical microwave approaches. This offers the ability to realise functions in microwave systems that are either very complex or, perhaps not even possible to achieve with existing RF approaches. Applications include front-end architectures with wide tunability combined with fibre-fed distributed antenna remoting in applications such as defence which require processing of broadband spatially distributed microwave signals  and in wireless-optical systems.
In this paper, we focus on recent methods in wideband and adaptive signal processing, which address the challenge of realising programmable microwave photonic phase shifters and true-time delay elements for phased array beamforming; ultra-wideband Hilbert transformers and quadrature filters; microwave photonic filters that can realise single passband, widely continuously tunable filtering, and switching operation; together with ultra-wideband microwave photonic mixers. Finally we present a new microwave photonic mixer structure, which is based on using the inherent frequency selectivity of the stimulated Brillouin scattering loss spectrum to suppress the carrier of a dual-phase modulated optical signal. Results demonstrate an extremely wide multi-decade bandwidth of 0.2 to 20 GHz together with a large conversion efficiency improvement comparison to the conventional dual-series Mach Zehnder modulator based microwave photonic mixer, while also enabling operation for antenna remoting.
2. Programmable microwave photonic elements for phased array beamforming
Optically controlled beamforming techniques for phased array antennas are of significant interest due to the key advantages that photonics can offer. These include a wide operating bandwidth, remote antenna feeding, and immunity to EMI. Programmable phase shifters are required for agile beamforming. However it is difficult to realise phase shifters that can operate over a very wide microwave frequency range using traditional electronic approaches. Techniques based on microwave photonics can offer a solution to this limitation.
Various structures to realise an RF phase shifting operation have been reported. RF phase shifters based on the slow light effect in semiconductor optical amplifiers (SOAs) have been reported , wherein the RF phase shift is controlled by the injection currents into the three cascaded SOAs, though this phase shifter showed a frequency dependent phase shift performance. Microwave photonic phase shifting using a single sideband (SSB) polarisation modulator and a polariser has been reported , where shifting the phase of an RF modulated optical signal at the polarisation modulator output is realised by adjusting the polarisation direction of the polariser in front of the photodetector. Slow and fast light effects in a tilted fibre Bragg grating has also been reported to realise the RF phase shifting operation . In this technique, an SSB modulated optical signal is sent to a tilted fibre Bragg grating with the optical carrier located at one cladding mode resonance of the spectral response to introduce a phase shift to the optical carrier. The beating between the phase-shifted optical carrier and the sideband generates a microwave signal with the phase shift from the optical carrier directly transferred to the generated microwave signal, though this approach has demonstrated a limited phase shift range of 280°. Another technique for a microwave photonic phase shifter that is implemented using a dual parallel Mach Zehnder modulator (DPMZM) followed by an optical bandpass filter is reported in . An optical bandpass filter is used to filter out one sideband of the double sideband suppressed carrier modulated optical signal in the lower arm of the DPMZM. The optical phase difference between the carrier, which passes through the upper arm of the DPMZM, and the sideband in the lower arm, is controlled by adjusting the modulator bias voltage. This optical phase difference is converted to an RF phase shift after photodetection, though this technique requires a sharp roll-off optical bandpass filter, and both the laser wavelength and the optical filter centre wavelength need to be precisely controlled.
A structure for a microwave photonic phase shifter that can achieve a full 360° phase shift with very little RF signal amplitude variation, and which can operate over a very wide frequency range is shown in Fig. 1. It is based on controlling the amplitude and phase of the optical carrier and the two RF modulation sidebands via the DC bias voltages to a dual-parallel Mach-Zehnder modulator . The two RF modulation sidebands, which have different amplitudes and phases, beat with the optical carrier at the photodetector to generate the output phase shifted RF signal. The amplitude and phase of this output RF signal depends on the amplitude and phase of the optical carrier as well as the differing amplitudes and phases of the two RF modulation sidebands. The two RF modulation sidebands that have different amplitudes and phase are set through the use of a DPMZM for the modulator unit. The DC bias voltages control the amplitude and phase of the optical carrier and the RF modulation sidebands. The structure is simple, only requiring a single laser, a single DPMZM unit that is commercially available, and a single photodetector. It has the advantage of having a very convenient control of the RF phase shift which is set by adjusting DC voltages. Regarding tuning speed, the inherent response time of the DPMZM is extremely fast being in the sub-ns range, so the tuning speed of the phase shifter is set by the response time of the DC power supply. Typical commercially available DC power supplies have a transient response time of less than 50 μs when the output current is changed from full load to half load, and since here the load current is negligible and only the voltage needs to be changed, the response time is expected to be even faster.
Figure 1 shows the dissimilar sidebands generated by the DPMZM when driven by a single frequency RF signal, and the output RF signal with a phase shift of θRF after photodetection. The idea is based on designing the amplitudes (Ac, ALSB and AUSB) and the phases (θc, θLSB and θUSB) of the optical carrier and the two RF modulation sidebands by controlling the modulator bias voltages (Vb1, Vb2 and Vb3), so that the output RF signal phase θRF can be shifted between 0° and 360° while the output RF signal amplitude ARF remains unchanged.
The measured phase and amplitude response of the phase shifter over a wide frequency range from 2 GHz to 16 GHz for different modulator bias voltages, are shown in Figs. 2(a) and 2(b). This demonstrates a flat phase response performance over a wideband 8:1 frequency range. The phase ripple standard deviation is less than 2° for all the measured phase responses in Fig. 2(a). Furthermore, Fig. 2(b) shows that there are less than 3 dB changes in the RF signal amplitude over the entire broadband frequency range during the 0° to 360° phase shift operation, which is one of the flattest responses reported for a wideband RF phase shifter.
Another approach for phase shifting based on the use of spatial light modulators is shown in Fig. 3 . Phase shifting is achieved by adjusting the relative optical phase of the carrier and the sideband within the full 0° to 360° range by means of a two-dimensional (2D) diffraction-based Fourier-domain optical processor (FD-OP) comprising an array of liquid crystal on silicon (LCoS) pixels , and this phase shift is directly translated to the RF signal after detection. Multiple wideband phase shifters are be realised within the single structure by processing the WDM wavelengths simultaneously. The LCoS pixels enable dynamic wavelength routing to the output ports, and optical to RF signal conversion with direct phase and amplitude translation control. The diffraction grating disperses and images different spectral components of the input modulated light onto a different portion of the LCoS horizontally while overlapping a large number of vertical pixels. Then specially calculated phase modulation patterns are applied to impress optical phase differences between the respective optical carrier and the sideband for each WDM channel, which are directly translated to the RF signal after detection. This beamforming network is highly flexible and reconfigurable since it is software programmable, and since the microwave phase shifters in the array are independent and have quasi-continuous phase control from 0 to 2π, arbitrary scanning beam angles can be realised. The tuning speed is limited by the update time of the LCoS system which is about 100 ms . Regarding the resolution, it can be noted that there are strong and ongoing driving forces to increase the LCoS resolution that come from important consumer electronics applications such as TV and projection systems, 3-D imaging, and defence. Significantly higher resolution has been reported for commercially available 2-D LCoS displays that can support an increased number of 1920x1080 pixels with a reduced pixel size down to 8 µm .
Experimental results are shown in Figs. 4(a)-4(d), which demonstrate four independent microwave phase shifters that operate simultaneously over a 10-20 GHz frequency range. Figure 4(a) shows a set of four phase shifters to obtain a scanning angle of 20° (with half wavelength radiating element spacing). Figure 4(b)-4(d) shows the flexibility of the structure for tuning the beam direction for scanning angles of −20°, 40° and −40°, which was achieved by programming the RF phase shifters. This approach can enable scalability to larger phased arrays due to the parallel processing capability of the concept.
As well as phase shifters, true-time-delay elements are critical elements in phased array antennas that are required to operate with wide instantaneous bandwidth. A technique to realise an array of multiple true-time-delay elements, which can be independently and continuously tuned, is shown in Fig. 5. It is based on a WDM parallel signal processing approach in conjunction with a Fourier-domain optical processor FD-OP . The WDM carriers are modulated by the input RF signal using an electro-optic modulator (EOM) with single-sideband with carrier (SSB + C) scheme. The modulated carriers are distributed to different locations on the LCoS array and the different true-time-delays and RF powers are controlled via programming of the diffraction-based Fourier-domain processor. Hence, true-time-delay RF modulated signals are obtained by programming the LCoS system to realise desired phase slopes. This is followed by a demultiplexer which routes the different wavelengths to individual photodetectors that output the RF signals. The technique features the ability to scale to realise a large number of true-time-delay lines while sharing a single optical processing device. It also features the ability to control the signal amplitude at the same time as controlling the true-time-delays.
Experimental results demonstrating a true-time-delay element with independent programmable tuning control, and minimal RF amplitude and group delay variation across a 4-20 GHz operation range are shown in Figs. 6(a) and 6(b). The LCoS system which was configured to provide a programmable optical field transfer function that had a linear optical phase response whose slope could be tuned, and the linear optical phase-shift is directly translated to the time delay of the RF signal. By varying the phase slope, different time delays are realised. Figure 6(a) shows the measured RF phase versus RF frequency for different phase slopes, which was achieved by programming the linear phase slope continuously by means of the LCoS system. Excellent high-linearity phase responses can be observed from the measurement results. The corresponding time delays are shown in Fig. 6(b). The tuning range is from −32 ps to + 32 ps. The flat response in Fig. 6(b) indicates a true RF time delay.
3. Ultra-wideband Hilbert transformer
Amongst the fundamental signal processing functions, the temporal Hilbert transform is an important function that is widely used in communication systems, radar, information processing, computing, signal analysis and measurement. The ideal Hilbert transform corresponds to a quadrature filter that operates over a very wideband frequency-independent range. However electronic implementations based on broadband 90° hybrid couplers are limited by the excessive amplitude and phase ripples that occur under broadband operation. Wideband microwave photonic Hilbert transformers exhibiting a wide operation bandwidth have been demonstrated recently using structures based on a fibre Bragg grating , integrated planar Bragg grating devices , and Fourier domain optical processors (FD-OP). The use of FD-OP enables a flexible optical wavelength allocation and programmability of the Hilbert transformer. Figure 7 shows a microwave photonic quadrature filter based on an all-optical Hilbert transformer  that uses an FD-OP. The principle is founded on the mapping of a photonic Hilbert transformer from the optical to the electrical domain, using spectral manipulation in the optical frequency domain by means of a programmable FD-OP. The input arbitrary RF signal modulates two optical wavelengths, namely λ1 and λ2 via the EOM, and is then launched into the FD-OP. The FD-OP comprises a two-dimensional LCoS array. This enables the spectral magnitude and phase response of the input light to be manipulated independently in the Fourier-domain, implementing an optical transfer function centred at 1550 nm wavelength with a bandwidth B of 100 GHz such that double-sideband modulated light at λ1 (set at 1550 nm) experiences a transfer function, corresponding to an optical Hilbert transform, as shown in Fig. 7, while simultaneously implementing an optical transfer function centred at 1551 nm with a bandwidth of B such that modulated light at λ2 (set at 1551 nm) undergoes a zero-phase all-pass optical filter response. This also shows the broadband generation of in-phase and quadrature (IQ) orthogonal components of an input microwave signal, which is important for vector modulation and signal processing, by directly translating the optical magnitude and phase response of an all-pass zero-phase filter and an all-optical Hilbert transform transfer function to the electrical domain.
Experimental results have demonstrated the realisation of an extremely wide operating bandwidth for a microwave quadrature filter with an amplitude imbalance of less than +/− 0.23 dB and a phase imbalance of less than +/−0.5° from 10 GHz to 20 GHz. The upper frequency limit is limited in practice by the modulation bandwidth of the EOM, and with the availability of wideband electro-optic devices, the system can be expected to operate in excess of 100 GHz.
4. Microwave photonic filters
4.1 Single passband microwave photonic filters
The ability to realise programmable and reconfigurable signal processors is important for adaptive applications. Moreover, it is essential to have a single passband response. However, conventional microwave photonic filters are limited by the presence of multiple harmonic passbands in their frequency response. This is a generic phenomenon that intrinsically occurs in all discrete time signal processors, and is a significant drawback because it restricts the processing frequency to a fraction of the free spectral range (FSR) in order to avoid spectral overlapping.
Various approaches to increase the filter FSR or to realise a single passband bandpass filter response have been reported including structures based on a silicon-on-insulator ring resonator  or phase shifted fibre Bragg gratings [19, 20]. However, these RF responses have limited programmability as only central frequency tuning was realised in the demonstrations.
A technique that can overcome the problem of multiple harmonic passbands in the frequency response is to use filtering based on optical mapping to the RF domain . Figure 8(a) shows a microwave photonic single passband filter based on using an electro-optical phase modulator and a tunable optical filter and only requires a single wavelength source and a single photodetector.
If the laser frequency ω0 is slightly displaced by Δω from at the central angular frequency of the optical filter ωc as shown in Fig. 8(b), (e.g. ωc = ω0 + Δω), there is a frequency offset between the optical filter and the modulated optical signal which leads to an asymmetrical suppression of the sidebands. Within the passband of the optical filter, the beating between the optical carrier with the optical sideband within the angular frequency region from ωc–B/2 to ω0 cancels with the beating between the carrier and the upper sideband within the angular frequency region from ω0 to ω0 + B/2–Δω at the photodetector, where B is the filter bandwidth. However the optical upper sideband signal within the angular frequency region from ω0 + B/2–Δω to ωc + B/2 remains, while the corresponding lower sideband is suppressed by the stopband of the optical filter. This results in the generation of a single passband RF response as only the beating between the remaining upper sideband with the optical carrier exists after the photodetection. The centre frequency of the RF filter is B/2, which is related to half of the bandwidth of the optical filter, and the bandwidth of the RF filter is 2Δω which corresponds to the frequency spacing between the optical carrier and the central frequency of the optical filter. In comparison to previous approaches [18–20], this approach has the capability of tuning the centre frequency and filter bandwidth independently. The RF filter centre frequency can be tuned by enlarging or reducing the optical filter bandwidth which can be implemented via tunable optical gratings . Moreover, the RF filter bandwidth can be adjusted simply by using a tunable laser to change the optical wavelength, which offers the potential advantage of fast tuning speed, or by altering the central frequency of the optical filter.
Many applications require microwave filters with high passband selectivity, which is difficult to achieve. The use of stimulated Brillouin scattering (SBS) in optical fibre is an attractive approach for implementing high-resolution microwave photonic filters. Following pioneering work in the application of SBS in microwave photonic signal processing , it has been shown that SBS can be used to obtain high Q-factor microwave photonic filters due to its inherent narrowband nature . Figure 9 shows the measured response of an SBS-based microwave photonic filter operating at a centre frequency of 10 GHz . A single passband response can be observed. The filter exhibits an extremely high resolution, having a −3 dB bandwidth of 21.25 MHz. In addition it has an excellent shape factor, having a −20 dB bandwidth of 170 MHz, and exhibits a high out-of-band rejection of 35 dB.
4.2 Widely and continuously tunable high-resolution filters
The ability to obtain wide continuous tunability together with very high-resolution bandpass filtering has been a long-standing problem that has been very difficult to achieve. Microwave photonic signal processing offers the means to solve this problem, and to realise such functions in microwave systems that are extremely challenging and may not even be possible in the RF domain. SBS-based microwave photonic filtering has enabled the realisation of an ultra-wide and continuous tuning range bandpass filter with continuous measured tuning range from 1 to 20 GHz  together with an extremely high Q of around 1000. The filter exhibits extremely high resolution with a −3 dB bandwidth of only 16 MHz, and a −20 dB bandwidth of 130 MHz. Moreover it has an excellent shape factor, having a −20 dB bandwidth of 170 MHz and exhibits a high out-of-band rejection of 35 dB. The filter realises single passband shape-invariant tuning over the entire 1-20 GHz tuning range.
The SBS principle can also be applied to realise a widely tunable narrow notch filter, and tunable microwave photonic notch filters have been realised demonstrating a continuous measured filter tuning range from 2 GHz to 20 GHz . The notch filter exhibits a very high resolution, having a −3 dB bandwidth of 82 MHz, and realises a deep notch with >40 dB suppression over the tuning range.
Other techniques for realising single passband RF filtering based on broadband optical sources have also been demonstrated recently, via either optical filtering  or non-sliced broadband optical source . An extension of  with the inclusion of a programmable optical spectrum processor and differential detector has been reported . By controlling the shape of the optical spectrum and the time delay of the Mach Zehnder interferometer in the filter structure a tunable, single-passband, flat-top and arbitrary shape bandpass filter response can be obtained. A principle for obtaining a single passband microwave photonic processor with wideband tunability, based on a dual input EOM that utilizes its specific optical configuration and the special π-shifted RF modulation on the optical signals entering and exiting it at the complementary ports, is shown in Fig. 10(a) . The broadband optical source is incident into a dual-input EOM with unbalanced input fibre lengths, which can be controlled by means of the optical variable delay line (VDL), from the optical source into the upper and lower arms of the EOM. The modulated signal is then launched into the linearly chirped fibre Bragg grating (LCFBG) though an optical circulator which applies a frequency-dependant group delay function. This topology consequently shifts the spectral curve of dispersion induced RF fading from the baseband location to high frequencies, where an equivalent single passband response is realised without the requirement of the conventional tap coefficients. The resulting filter response is baseband-suppressed and can be continuously and widely tuned to arbitrary high RF frequencies without varying the filter shape.
The amplitude response is the frequency-shifted version of the baseband RF induced fading curve, as illustrated by the single passband response in Fig. 10(b), where the centre frequency of the RF fading is moved from the baseband to the angular frequency of ωs which is controlled by the length difference introduced by the VDL and the group delay slope of the LCFBG. The tunability of the response can be achieved by adjusting the length of the VDL.
Figure 11 demonstrates the wide and continuous tunability of the photonic signal processor as the VDL was adjusted to introduce various optical delays between the two interferometric arms of the 50/50 optical coupler before entering the dual-input EOM, corresponding to centre frequencies from 2 GHz to 18 GHz with an increment of 2 GHz. This shows that the single passband response filter is tuned continuously over the entire measurement range (20 GHz). It can be noted that the passband response has been shape-invariantly tuned over the range without any distortion.
Techniques for obtaining extremely fast tuning of the passband frequency of microwave photonic filters have been reported in [30, 31]. These approaches are based on controlling the relative delay between a synchronized pair of combs. In  an optical comb generated by a strongly phase modulated continuous wave light is passed through a line-by-line pulse shaper for controlling the intensity of the comb lines. The comb lines are then intensity modulated by a SSB modulator and pass through a dispersion compensating fibre, which introduces a time delay to each comb line, before photodetection. In  the optical combs are generated by periodic intensity and phase modulation of a continuous wave (CW) laser and are then passed through a SSB modulator driven by the RF input signal which then propagates through a dispersive fibre so the comb lines arrive at the photodetector with different delays and hence function as independent taps. Because the two comb generators are driven at the same RF frequency, the relative delay of the generated optical trains is governed by the relative phase shift of the respective RF drive signals, and this can be electronically tuned rapidly using an analogue RF phase shifter. Using this approach, very fast tuning of the passband frequency has been demonstrated with:40 ns speed by controlling the optical delay between combs .
Emerging future directions include the trend towards integration. InP-based photonic integrated circuits for microwave photonics have been reported in , including unit cells for microwave photonic lattice filters whose response can be varied using semiconductor optical amplifiers. Integrated microwave photonic filters on a hybrid silicon platform based on micro-ring resonators have also been reported . Another technique for realising integrated microwave photonic filtering is by using silicon photonic technology together with coherent detection . The filtering operation relies on the use of a tunable CMOS optical filter. In addition to the filtering operation, the structure can also realise the frequency downconversion operation by using optical heterodyne detection with an injection locked photonic LO. As with other silicon photonic devices, the temperature needs to be precisely controlled in order for this filter to have a stable performance. Microwave photonic filters that are implemented on a compact chalcogenide photonic chip have also been reported , which are operated based on stimulated Brillouin scattering. These filters have shown a sharp passband of around 23 MHz, a shape factor of 2 to 3.5, and a tuning range of 2 to 12 GHz, however the filter stopband rejection level was only around 20 dB. Recently, photonic crystal waveguides whose dispersion profile can be tailored, have been used to implement tunable microwave photonic filters based on a 1.5-mm-long GaInP/GaAs highly dispersive photonic crystal waveguide delay line . A controllable delay with a total insertion loss of around 10 dB when the delay is below 70 ps has been demonstrated. Both 2-tap and 4-tap microwave photonic filters have been realised using multiple WDM laser sources with the photonic crystal chip. Future work in this area aims to obtain fully integrated devices that can realise more complex and elaborate filter functions.
4.3 Switchable microwave photonic filters
Multi-function applications require not just a fixed transfer function but the ability to switch the different filtering functions e.g. between a bandpass filter and a notch filter to provide both channel selection and channel rejection. A structure that enables the realisation of a switchable microwave photonic filter that can be switched between a bandpass filter and a stopband notch filter response, using simple and rapid control is shown in Fig. 12(a) . It is based on a SBS technique in conjunction with a dual drive Mach-Zehnder modulator (DDMZM) that processes the sidebands of the RF modulated signal. Switching of the filter function is simply and conveniently obtained by changing the DC bias to the DDMZM. In addition, the centre frequency of the switchable filter can be tuned over a wide frequency range.
The pump light is modulated at RF frequency fp by a low-biased intensity modulator (IM) in the lower branch which generates a double-sideband suppressed-carrier signal, and which is used to tune the filter frequency by changing the positions of the SBS gain and loss. The filter switching function is achieved by changing the DC bias voltage VDC of the DDMZM to control the relative phase differences between the carrier and the sidebands. The SBS process occurs in the fibre between the RF modulated signal and the pump signal. Each sideband of the pump signal introduces both SBS gain and SBS loss spectra at frequency fB away from the sideband, where fB is the Brillouin frequency shift.
In order to realise bandpass filtering, the bias is selected so that the DDMZM operates as a phase modulator. Only the RF signal with the frequency of fRF = fp-fB undergoes the SBS effect which consequently breaks the amplitude equality between the sidebands since the lower sideband of the RF modulated signal is significantly attenuated while the upper sideband is amplified, as shown in the right-hand side of Fig. 12(b), and this creates a bandpass response. In order to realise notch filtering, we select the bias so that the phase difference between the upper sideband and the carrier is π/4 while the phase difference between the lower sideband and carrier is 3π/4, and in addition there is a large amplitude imbalance between the sidebands, as shown in Fig. 12(b). Hence, an RF signal is obtained at the photodetector output at all frequencies except when the input RF frequency is fRF = fp-fB, and the amplitudes of the two sidebands are equalised by properly choosing the SBS pump power, as shown in the right-hand side of Fig. 12(b), when the beatings between the carrier and the two sidebands are out-of-phase and fully cancel, thus creating a notch. Consequently, a switchable microwave photonic filter at the centre frequency of fp-fB is realised, in which the switching function is easily accomplished by changing the DC bias of the DDMZM. The frequency where the switched filtering occurs can also be tuned by changing the drive frequency fp.
Experimental results are shown in Figs. 13(a) and 13(b) which demonstrate the ability of this structure to switch between a high-resolution bandpass filter and a high-resolution notch filter at each frequency, with Q values around 400 to 500, and the ability to operate over a wide frequency range from 2 to 20 GHz. Tuning of the filter was obtained by changing the pump frequency fp. Switching between high-resolution bandpass and notch filtering with 3-dB widths of 30-40 MHz, at all frequencies from 2 to 20 GHz with shape-invariant response is demonstrated. The Brillouin process involves interesting interactions and further investigation of the mechanism is useful to elucidate and optimise the characteristics of SBS for microwave photonic applications.
5. Ultra-wideband microwave photonic mixers
Microwave frequency conversion is a key function that is essential in many optical-wireless system applications. Advanced communication and radar systems need to operate over multiple passbands simultaneously. Photonic microwave mixers bring advantages of extremely wide bandwidth of operation, near infinite isolation between the RF and the LO ports, and EMI immunity, which are unique and fundamental features of photonics. They are also attractive for processing microwave and RF signals directly within the optical fibre transport system.
Various approaches to realise frequency conversion of microwave signals in the optical domain have been reported recently. The technique based on using a coherent RF-to-bits photonic receiver  can obtain low conversion loss and a high linearity performance in the millimetre wave frequency band. It relies on the use of a micro-optic 90° hybrid coupler, which by interfering in the two arms of the fibre interferometer, creates the downconverted in-phase and quadrature-phase photocurrents. However, a feedback loop is required to stabilise the optical phase difference of the RF and LO modulated optical signals by means of adjusting a long-throw fibre stretcher. An approach based on a Mach Zehnder modulator (MZM) in series with a linearised DPMZM has been reported  to obtain a high spurious free dynamic range (SFDR) frequency converter. The local oscillator (LO) is applied to a low-biased MZM and the RF signal is applied to the DPMZM. By optimising the bias voltages of the DPMZM the third order nonlinear components are suppressed to obtain an increased SFDR. An interesting approach for frequency conversion based on the use of quadratic optical phase modulators has been reported . Due to the quadratic nonlinearity of the phase modulators, the mixing product between the RF and LO is encoded on the optical phase. The optical phase is then recovered through an optical phase locked loop linear phase demodulator. A conversion loss of 16 dB and a SFDR of 116 dBHz2/3 have been measured using the shot noise limited noise floor. Again an optical phase stabilisation feedback circuit with a fibre line stretcher is required to obtain a stable output.
The most common microwave photonic mixer structure is based on a series connection of two electro-optic intensity modulators , where the RF signal and the LO are applied to the two separate optical intensity modulators. However, this suffers from high losses and very low conversion efficiency, defined as the ratio of the output IF signal power to the input RF signal power. This is a significant limitation for microwave frequency conversion. This problem cannot be overcome by increasing the optical power because of the limited optical power handling ability of the photodetector. In fact the average optical power of this downconverter is dominated by the optical carrier, which is undesirable since it does not contribute to the downconversion process that is actually obtained by the beating of the RF signal and the LO sidebands at the photodetector, as shown in Fig. 14.
The basic approach to solve this problem is to suppress the optical carrier so as to enable higher RF signal and LO sidebands to be incident onto the photodetector, thus resulting in a higher output IF signal power and consequently higher conversion efficiency. A technique to implement this has been previously reported using an optical filter to suppress the optical carrier at the output of a series connection of two phase modulators . However, this introduces critical requirements on fixing the centre wavelength of the optical filter, which also must be insensitive to environmental changes in order to realise large carrier suppression without affecting the RF information signal and the LO, and it also requires an ultra-narrow optical filter bandwidth, making this approach difficult.
A microwave photonic mixer structure that overcomes this problem is shown in Fig. 15. It is based on an integrated DPMZM and optical phase shifter configuration . This structure features the advantages of high conversion efficiency, robust operation, and ability to function over a very wide frequency range. The integrated DPMZM comprises four optical phase modulators and an optical phase shifter. The upper pair of the optical phase modulators forms a MZ modulator in a push-pull configuration, to which the LO together with a DC bias voltage are applied. The lower pair of the optical phase modulators also forms a MZ modulator in a push-pull configuration, to which the RF information signal together with a DC bias voltage is applied. In addition, this branch contains an optical phase shifter. The output of this parallel connected MZ structure is combined coherently before being detected by the photodetector.
To maximise the conversion efficiency of the microwave photonic mixer, the integrated DPMZM based photonic mixer is designed so that the carrier is suppressed, and the IF signal is generated by the beating between the RF information signal sidebands and the LO sidebands only at the photodetector, as shown in Fig. 16. The upper MZ modulator generates the two LO sidebands in Fig. 16, and the lower MZ modulator generates the two RF sidebands in Fig. 16, with the optical carrier being eliminated to reduce the average output optical power into the photodetector. A high power IF signal is obtained by using an erbium-doped fibre amplifier (EDFA) to amplify the RF and LO sidebands. The optical filter after the EDFA is simply used to reduce the amplified spontaneous emission (ASE) noise generated by the EDFA. It can be noted that the EDFA and the optical filter can be eliminated by using a high power laser source and an integrated DPMZM with high optical power handling ability.
Figure 17 shows the measured conversion efficiency of the integrated DPMZM based photonic mixer as a function of input RF signal frequency. The integrated DPMZM had a 3-dB bandwidth of 16.7 GHz. The LO frequency was adjusted according to the RF signal frequency to keep the downconverted IF signal frequency constant at 80 MHz. A high LO suppression of 45.5 dB was measured at the output. The experimental results demonstrated a significant improvement of 23.7 dB in the conversion efficiency compared to the conventional dual-series Mach Zehnder intensity modulator based photonic mixer  for the same optical power into the photodetector. It can be seen from the Fig. 17 that the photonic mixer response between 2 and 16 GHz is flat with around 1 dB variation.
While this DPMZM approach can suppress the optical carrier, it has the disadvantage that the downconversion operation is performed in a single unit, which does not permit the modulators driven by the RF signal and the LO to be located in different locations where for example the RF is at a remote antenna and the LO is at the base station. This is required in important applications such as for antenna remoting in wireless-optical systems, and in defence receiver applications  that require sensor RF antennas to be located at physically disparate locations around the host platform for direction finding or obscuration avoidance while the LO is placed at an environmentally controlled base station.
To solve this problem, we present a new microwave photonic mixer. It is based on using the inherent frequency selectivity of the SBS loss spectrum to suppress the carrier of a dual-phase modulated optical signal. This not only increases the conversion efficiency to enable high conversion efficiency mixing operation to be obtained over a wide frequency range, but at the same time it also enables antenna remoting for the RF input location to be realised.
The structure of the new microwave photonic mixer is shown in Fig. 18. The laser light is split into two paths via an optical coupler. The light travelling in the upper path is phase modulated by the RF information signal and is then phase modulated again by the LO. This generates an RF signal and LO sideband on each side of the optical carrier with 180° phase difference, which is launched into an optical fibre through an optical isolator. The lower arm of the structure shown in Fig. 18 comprises an optical frequency shifter. The frequency shifter, implemented by a DPMZM driven by a single frequency tone, down shifts the laser frequency by the optical fibre SBS frequency. This frequency shifted light is amplified and is injected into the other end of the optical fibre via an optical circulator. It travels in a counter-propagating direction to the forward propagating phase modulated optical signal. This introduces a SBS loss spectrum in the Stokes wave at the optical carrier frequency and consequently suppresses the carrier of the phase modulated optical signal. The optical amplifier after the frequency shifter is used to control the pump power to ensure large suppression of the optical carrier to be below the LO sidebands. The optical amplifier after the circulator amplifies the RF signal and LO sideband amplitudes before beating at the photodetector to generate a high-power IF signal with the frequency ωIF = ωRF-ωLO as shown in Fig. 19. Hence a high conversion efficiency mixing operation is obtained.
Figure 20 shows the measured SBS based mixer conversion efficiency as a function of the input RF signal frequency, with the LO frequency being adjusted to maintain a constant output IF signal frequency of 160 MHz. The experimental results demonstrate an extremely wide bandwidth operation of 0.2 to 20 GHz. In addition, measurements showed that the SBS based microwave photonic mixer has a large conversion efficiency improvement of over 26 dB in comparison to the conventional dual-series MZM based microwave photonic mixer  operating under the same conditions.
Photonic signal processing offers the advantages of large time-bandwidth capabilities to overcome inherent electronic limitations. It provides attractive features for RF pre-processing with EMI immunity in fibre-fed systems, and enables in-fibre processors with connectivity and in-built signal conditioning. Recent methods in wideband and adaptive signal processing, which address the challenge of realising programmable microwave photonic phase shifters and true-time delay elements for phased array beamforming; ultra-wideband Hilbert transformers and quadrature filters; microwave photonic filters that can realise single passband, widely continuously tunable filtering, and switching operation; together with ultra-wideband microwave photonic mixers, have been described. Also, a new microwave photonic mixer structure has been presented, which is based on using the inherent frequency selectivity of the SBS loss spectrum to suppress the carrier of a dual-phase modulated optical signal. Experimental results have demonstrated an extremely wide bandwidth operation of 0.2 to 20 GHz together with a conversion efficiency improvement of over 26 dB in comparison to the conventional dual-series MZM based microwave photonic mixer, while also enabling antenna remoting for the RF input location to be realised. These processors provide new capabilities for the realisation of high-performance and high-resolution signal processing.
This work was supported by the Australian Research Council. Thanks are extended to T. X. Huang and W. Zhang for their valuable contributions to this work.
References and links
1. R. A. Minasian, “Photonic signal processing of microwave signals,” IEEE Trans. Microw. Theory Tech. 54(2), 832–846 (2006). [CrossRef]
2. J. Capmany, B. Ortega, and D. Pastor, “A tutorial on microwave photonic filters,” J. Lightwave Technol. 24(1), 201–229 (2006). [CrossRef]
3. J. P. Yao, “Microwave photonics,” J. Lightwave Technol. 27(3), 314–335 (2009). [CrossRef]
4. D. Dolfi, “New trends in optoelectronics for radar, E.W. and communication systems,” IEEE International Topical Meeting on Microwave Photonics (MWP 2011) 7 (2011).
5. W. Xue, S. Sales, J. Capmany, and J. Mørk, “Wideband 360° microwave photonic phase shifter based on slow light in semiconductor optical amplifiers,” Opt. Express 18(6), 6156–6163 (2010). [CrossRef] [PubMed]
6. S. Pan and Y. Zhang, “Tunable and wideband microwave photonic phase shifter based on a single-sideband polarization modulator and a polarizer,” Opt. Lett. 37(21), 4483–4485 (2012). [CrossRef] [PubMed]
8. J. Shen, G. Wu, W. Zou, and J. Chen, “A photonic RF phase shifter based on a dual-parallel Mach-Zehnder modulator and an optical filter,” App. Phy. Express 5(7), 072502 (2012). [CrossRef]
9. E. H. W. Chan, W. Zhang, and R. A. Minasian, “Photonic RF phase shifter based on optical carrier and RF modulation sidebands amplitude and phase control,” J. Lightwave Technol. 30(23), 3672–3678 (2012). [CrossRef]
10. X. Yi, T. X. H. Huang, and R. A. Minasian, “Photonic beamforming based on programmable phase shifters with amplitude and phase control,” IEEE Photon. Technol. Lett. 23(18), 1286–1288 (2011). [CrossRef]
11. M. A. F. Roelens, S. Frisken, J. A. Bolger, D. Abakoumov, G. Baxter, S. Poole, and B. J. Eggleton, “Dispersion trimming in a reconfigurable wavelength selective switch,” J. Lightwave Technol. 26(1), 73–78 (2008). [CrossRef]
12. J. Schröder, O. Brasier, T. D. Vo, M. A. F. Roelens, S. Frisken, and B. J. Eggleton, “Simultaneous multi-channel OSNR monitoring with a wavelength selective switch,” Opt. Express 18(21), 22299–22304 (2010). [CrossRef] [PubMed]
13. F. Yaras, H. Kang, and L. Onural, “State of the art in holographic displays: A survey,” J. Disp. Technol. 6(10), 443–454 (2010). [CrossRef]
14. X. Yi, L. Li, T. X. H. Huang, and R. A. Minasian, “Programmable multiple true-time-delay elements based on a Fourier-domain optical processor,” Opt. Lett. 37(4), 608–610 (2012). [CrossRef] [PubMed]
15. M. Li and J. Yao, “Experimental demonstration of a wideband photonic temporal Hilbert transformer based on a single fiber Bragg grating,” IEEE Photon. Technol. Lett. 22(21), 1559–1561 (2010). [CrossRef]
16. C. Sima, J. C. Gates, H. L. Rogers, P. L. Mennea, C. Holmes, M. N. Zervas, and P. G. R. Smith, “Phase controlled integrated interferometric single-sideband filter based on planar Bragg gratings implementing photonic Hilbert transform,” Opt. Lett. 38(5), 727–729 (2013). [CrossRef] [PubMed]
17. T. X. H. Huang, X. Yi, and R. A. Minasian, “Microwave photonic quadrature filter based on an all-optical programmable Hilbert transformer,” Opt. Lett. 36(22), 4440–4442 (2011). [CrossRef] [PubMed]
18. J. Palaci, G. E. Villanueva, J. V. Galan, J. Marti, and B. Vidal, “Single bandpass photonic microwave filter based on a notch ring resonator,” IEEE Photon. Technol. Lett. 22(17), 1276–1278 (2010). [CrossRef]
19. J. Palaci, P. Perez-Millan, G. E. Villanueva, J. L. Cruz, M. V. Andres, J. Marti, and B. Vidal, “Tunable photonic microwave filter with single bandpass based on a phase-shifted fiber Bragg grating,” IEEE Photon. Technol. Lett. 22(19), 1467–1469 (2010). [CrossRef]
20. W. Li, M. Li, and J. P. Yao, “A narrow-passband and frequency-tunable micro-wave photonic filter based on phase-modulation to intensity-modulation conversion using a phase-shifted fiber Bragg grating,” IEEE Trans. Microw. Theory Tech. 60(5), 1287–1296 (2012). [CrossRef]
22. A. Loayssa, D. Benito, and M. José Garde, “Applications of optical carrier Brillouin processing to microwave photonics,” Opt. Fiber Technol. 8(1), 24–42 (2002). [CrossRef]
23. B. Vidal, T. Mengual, and J. Marti, “Photonic microwave filter with single bandpass response based on Brillouin processing and SSB-SC,” IEEE International Topical Meeting on Microwave Photonics (MWP2009) 1–4 (2009).
24. W. Zhang and R. A. Minasian, “Widely tunable single-passband microwave photonic filter based on stimulated Brillouin scattering,” IEEE Photon. Technol. Lett. 23(23), 1775–1777 (2011). [CrossRef]
25. W. Zhang and R. A. Minasian, “Ultra-wide tunable microwave photonic notch filter based on stimulated Brillouin scattering,” IEEE Photon. Technol. Lett. 24(14), 1182–1184 (2012). [CrossRef]
26. M. Bolea, J. Mora, L. R. Chen, and J. Capmany, “Highly chirped reconfigurable microwave photonic filter,” IEEE Photon. Technol. Lett. 23(17), 1192–1194 (2011). [CrossRef]
27. X. Xue, X. Zheng, H. Zhang, and B. Zhou, “Widely tunable single-bandpass microwave photonic filter employing a non-sliced broadband optical source,” Opt. Express 19(19), 18423–18429 (2011). [CrossRef] [PubMed]
28. X. Xue, X. Zheng, H. Zhang, and B. Zhou, “Highly reconfigurable microwave photonic single-bandpass filter with complex continuous-time impulse responses,” Opt. Express 20(24), 26929–26934 (2012). [CrossRef] [PubMed]
29. L. Li, X. Yi, T. X. H. Huang, and R. A. Minasian, “Shifted dispersion-induced radio-frequency fading in microwave photonic filters using a dual-input Mach-Zehnder electro-optic modulator,” Opt. Lett. 38(7), 1164–1166 (2013). [CrossRef] [PubMed]
30. E. Hamidi, D. E. Leaird, and A. M. Weiner, “Tunable programmable microwave photonic filters based on an optical frequency comb,” IEEE Trans. Microw. Theory Tech. 58(11), 3269–3278 (2010). [CrossRef]
31. V. R. Supradeepa, C. M. Long, R. Wu, F. Ferdous, E. Hamidi, D. E. Leaird, and A. M. Weiner, “Comb-based radiofrequency photonic filters with rapid tunability and high selectivity,” Nat. Photonics 6(3), 186–194 (2012). [CrossRef]
32. L. A. Coldren, “Photonic integrated circuits for microwave photonics,” IEEE International Topical Meeting on Microwave Photonics (MWP 2010) 1–4 (2010). [CrossRef]
33. H. W. Chen, A. W. Fang, J. D. Peters, Z. Wang, J. Bovington, D. Liang, and J. E. Bowers, “Integrated microwave photonic filter on a hybrid silicon platform,” IEEE Trans. Microw. Theory Tech. 58(11), 3213–3219 (2010). [CrossRef]
34. K. Y. Tu, M. S. Rasras, D. M. Gill, S. S. Patel, Y. K. Chen, A. E. White, A. Pomerene, D. Carothers, J. Beattie, M. Beals, J. Michel, and L. C. Kimerling, “Silicon RF-photonic filter and down-converter,” J. Lightwave Technol. 28(20), 3019–3028 (2010). [CrossRef]
35. A. Byrnes, R. Pant, E. Li, D. Y. Choi, C. G. Poulton, S. Fan, S. Madden, B. Luther-Davies, and B. J. Eggleton, “Photonic chip based tunable and reconfigurable narrowband microwave photonic filter using stimulated Brillouin scattering,” Opt. Express 20(17), 18836–18845 (2012). [CrossRef] [PubMed]
36. J. Sancho, J. Bourderionnet, J. Lloret, S. Combrié, I. Gasulla, S. Xavier, S. Sales, P. Colman, G. Lehoucq, D. Dolfi, J. Capmany, and A. De Rossi, “Integrable microwave filter based on a photonic crystal delay line,” Nat. Commun.1–9 (2012). [CrossRef]
37. W. Zhang and R. A. Minasian, “Switchable and tunable microwave photonic Brillouin-based filter,” IEEE Photonics J. 4(5), 1443–1455 (2012). [CrossRef]
38. S. R. O’Connor, M. C. Gross, M. L. Dennis, and T. R. Clark, Jr., “Experimental demonstration of RF photonic downconversion from 4-40 GHz,” IEEE International Topical Meeting on Microwave Photonics (MWP 2009) 1–3 (2009).
39. S. Li, X. Zheng, H. Zhang, and B. Zhou, “Highly linear millimetre-wave over fiber transmitter will subcarrier upconversion,” Conference on Lasers and Electro-optics (CLEO 2011) 1–2 (2011). [CrossRef]
40. Y. Li, R. Wang, J. S. Klamkin, L. A. Johansson, P. R. Herczfeld, and J. E. Bowers, “Quadratic electrooptic effect for frequency down-conversion,” IEEE Trans. Microw. Theory Tech. 58(3), 665–673 (2010). [CrossRef]
41. G. K. Gopalakrishnan, W. K. Burns, and C. H. Bulmer, “Microwave-optical mixing in LiNbO3 modulators,” IEEE Trans. Microw. Theory Tech. 41(12), 2383–2391 (1993). [CrossRef]
43. E. H. W. Chan and R. A. Minasian, “Microwave photonic downconverter with high conversion efficiency,” J. Lightwave Technol. 30(23), 3580–3585 (2012). [CrossRef]
44. M. E. Manka, “Microwave photonics electronic warfare technologies for Australian defence,” IEEE International Topical Meeting on Microwave Photonics (MWP 2008) 1–2 (2009). [CrossRef]