Generation of orbital angular momentum multiplexing millimeter waves based on a circular traveling wave antenna

Rui Wang; Manting Wang; Youfei Zhang; Youfei Zhang; Youfei Zhang; Dashuang Liao; Dashuang Liao; Dashuang Liao; Liqiao Jing; Liqiao Jing

doi:10.1364/OE.483629

1. Introduction

The exceptional growth of the data traffic will exhaust the available capability for wireless communications in the future due to the increasing demands for portable multimedia streaming [1,2], augmented reality [3,4], vehicular networks [5–7], cloud services [8,9] and so on. To attain a higher data rate (>10∼100 Gbps), the carrier frequency must be pushed towards the upper millimeter wave for greater channel capacities. In particular, the large available and unlicensed spectrum at 60 GHz frequency can reach several Gbps [10]. Besides increasing the carrying frequency, the multiplexing technique is another efficient method to improve the channel capacity as it can convey multiple channels of the information at same frequency. Nowadays, time/space/frequency division multiplexing techniques have been widely used in wireless communications [11]. In addition to those multiplexing techniques, OAM-based mode division multiplexing provides another freedom and has attached extensive attention in academic and industry interest due to the inherent mutual orthogonality among each orbital angular momentum mode [12,13]. OAM antenna lies at the heart in the OAM-based wireless communication systems to transmit (Tx) and receive (Rx) a vortex beam carrying modulated OAM channel. Conventional methods for OAM generation include spiral phase plates [14], twisted parabolic reflectors [15], metasurfaces [16–20], phased array antennas [21], and dielectric resonators [22]. Moreover, the OAM beams can be reconfigured by introducing PIN diodes [23,24] or mechanical deformation [25]. The aforementioned OAM generation methods can only generate a single OAM mode, but multiplexing OAM modes with a shared aperture is a challenge especially in millimeter wave wireless communications.

Various approaches have been investigated to achieve OAM multiplexing. Two eigenmodes with 90° phase difference are excited in a circular cavity to generate OAM, whose topological charge depends on the feeding phase difference [26,27]. Based on this mechanism, many OAM multiplexing antennas are designed such as circular patch antenna [28], spoof localized surface plasmons [29], traveling wave ring resonators [30] and cylindrical dielectric resonator antennas [31]. The metasurface has emerged as a powerful platform to manipulate wave packets [32,33]. A holograph-inspired flat lens antenna is capable of simultaneously transmitting two independent coaxially propagating OAM beams [34]. The antenna array [35] is also efficient to form OAM multiplexing beams using different feeding networks, including dividing networks [36], Bulter matrix [37], Rotman lenses [38] and so on. However, the antennas mentioned above are not compact and antennas feed is far away from the array, making the system bulky and difficult to integrate into millimeter wave communication systems.

In this paper, we theoretically and experimentally demonstrated a cost-effective scheme for generating OAM waves based on the traveling wave circular structure. Multiplexing OAM modes can be generated and transmitted coaxially at 60 GHz using gradient waveguide ports, which are capable of integrating in the millimeter wave wireless communication systems. Then, we model the OAM antenna as a magnetic type circular loop current and derive the radiation fields. A prototype with OAM states l = ±3 carried by the z polarization and l = ±2 in the x and y polarizations at 60 GHz is fabricated and measured. The measured near-field distributions and far-field radiation patterns show excellent agreement with the simulated results. Further, a quad-OAM-mode multiplexing antenna is proposed at the operating frequency 60 GHz, which is composed of two concentrically stacked traveling wave circular radiators for generation of OAM with topological charges l = ±3 and l = ±5 in z polarization and l = ±2 and l = ±4 in x and y polarizations, respectively. Such a designer strategy provides a new approach for OAM-based multiplexing.

2. Results

Figure 1(a) depicts the configuration of the OAM circular traveling wave antenna, which is excited by gradient waveguide ports. The antenna will generate vortex beams with opposite topological charges by exciting different waveguide ports. Figure 1(b) shows the mathematical model of a circular loop with a current distribution of , in which ${\varphi _m}$ is the azimuthal angle, l is termed as the phase distribution parameter or can be called topological charge. The far field radiation can be generated by the M discrete azimuthal direction currents distributed along the circular loop with the radius a. The magnetic vector A is written as:

(1)$${\mathbf A} = \displaystyle{\mu \over {4\pi }}\oint_S {J({\mathbf r}^{\prime})} \displaystyle{{e^{-jk}\left| {{\mathbf r}-{\mathbf r}^{\prime}} \right|} \over {\left| {{\mathbf r}-{\mathbf r}^{\prime}} \right|}}ds^{\prime}$$

where ${\mathbf J}({{\mathbf r}^{\prime}}) = {\mathbf j} \cdot {e^{ - jl{\varphi _m}}} \cdot \delta ({{\mathbf r}^{\prime}} - {\mathbf r}_m^{\prime})$ is corresponding discrete current, r denotes the observation point, ${{\mathbf r}^{\prime}}$ represents the source point, ${\varphi _m} = \frac{{2\pi }}{M}m$; m = 0, 1, …, M−1, the m-th current is ${j} = |{j} |(\hat{{x}}\sin {\varphi _m} - \hat{{y}}\cos {\varphi _m})$ located at the position ${\mathbf r}_m^{\prime} = a(\hat{{\mathbf x}}\sin {\varphi _m} + \hat{{\mathbf y}}\cos {\varphi _m})$, the $\hat{{\mathbf x}}$ and $\hat{{\mathbf y}}$ is the unit vector along the x and y directions, respectively.

Fig. 1. (a) Schematic of the millimeter wave orbital angular momentum antenna. (b) Mathematical model of a circular traveling-wave antenna.

Download Full Size | PDF

With the far field condition, $|{{\mathbf r} - {\mathbf r}_m^{\prime}} |\approx r - {\mathbf r} \cdot {\mathbf r}_m^{\prime} \approx r$ and Euler’s formula, when l < 0, ignore high-order Bessel functions ${J_{l - 1}}(ka\sin \theta )$, Eq. (1) can be reduced to

(2)$$\begin{aligned} {\mathbf A} &= \frac{{\mu |{\mathbf j} |}}{{4\pi }}\frac{{{e^{ - jkr}}}}{r}{e^{ - jn\phi }}[ - \frac{{\hat{{\mathbf x}}}}{{2j}}M{( - j)^{l + 1}}{( - 1)^{l + 1}}{J_{l + 1}}(ka\sin \theta ){e^{ - j(1 + l)\phi }}\\ &\quad - \frac{{\hat{{\mathbf y}}}}{{2j}}M{( - j)^{l + 1}}{( - 1)^{l + 1}}{J_{l + 1}}(ka\sin \theta ){e^{ - j(1 + l)\phi }}] \end{aligned}$$

when l > 0, ignore high-order Bessel functions ${J_{l + 1}}(ka\sin \theta )$, Eq. (1) can be simplified as

(3)$$\begin{aligned} {\mathbf A} &= \frac{{\mu |{\mathbf j} |}}{{4\pi }}\frac{{{e^{ - jkr}}}}{r}{e^{ - jn\phi }}[\frac{{\hat{{\mathbf x}}}}{{2j}}M{( - j)^{l - 1}}{( - 1)^{l - 1}}{J_{l - 1}}(ka\sin \theta ){e^{ - j(l - 1)\phi }}\\ &\quad - \frac{{\hat{{\mathbf y}}}}{{2j}}M{( - j)^{l - 1}}{( - 1)^{l - 1}}{J_{l - 1}}(ka\sin \theta ){e^{ - j(l - 1)\phi }}] \end{aligned}$$

The electric field in the far-field region is calculated as:

(4)$${\mathbf E} ={-} j\omega {\mathbf A}$$

It can be seen that both the x- and y- components of the far field carry azimuth dependent phase, when l = ±3 and ±5, x- and y- components generate OAM with topological charges of ±2 and ±4, respectively.

Considering the OAM is observed in the transverse plane perpendicular to the z-axis in the paraxial region, the z-component in the far field is very small. However, the observed plane which is not far enough from the antenna, the phase term $|{{\mathbf r} - {\mathbf r}_m^{\prime}} |$ in (1) should be approximated with higher accuracy than far field condition that used for the analysis of the x- and y- components,

(5)$${{\mathbf r} - {\mathbf r}_m^{\prime}} |\approx r - {\mathbf r} \cdot {\mathbf r}_m^{\prime} = r - a\sin \theta \cos (\phi - {\varphi _m})$$

Thus, the magnetic vector A can be expressed as.

(6)$$\begin{aligned} {\mathbf A} &= \frac{{\mu a|{\mathbf j} |}}{{4\pi }}\frac{{{e^{ - jkr}}}}{r}\int_0^{2\pi } {(\hat{{\mathbf x}}\sin {\varphi _m} - \hat{{\mathbf y}}\cos {\varphi _m})} \cdot {e^{ - jl{\varphi _m}}} \cdot {e^{jka\sin \theta \cos (\phi - {\varphi _m})}}d{\varphi _m}\\ &={-} \frac{{{j^{ - l}}\mu a|{\mathbf j} |}}{4}\frac{{{e^{ - jkr}}}}{r} \cdot {e^{ - jl{\varphi _m}}}[(\hat{{\mathbf r}}\sin \theta + \hat{\boldsymbol{\mathrm{\theta}}}\cos \theta - \hat{\boldsymbol{\mathrm{\varphi}}}j){J_{l - 1}}(ka\sin \theta )\\ &\quad + (\hat{{\mathbf r}}\sin \theta + \hat{\boldsymbol{\mathrm{\theta}}}\cos \theta + \hat{\boldsymbol{\mathrm{\varphi}}}j){J_{l + 1}}(ka\sin \theta )] \end{aligned}$$

The electric and magnetic field can be obtained based on the vector potential,

(7)$${\mathbf H} = \frac{1}{\mu }\nabla \times {A}$$

(8)$${\mathbf E} = \frac{1}{{j\omega \varepsilon }}\nabla \times {\mathbf H}$$

the z-component is obtained as,

(9)$$\begin{array}{c} {{\mathbf E}_z} ={-} \frac{{{j^{ - l}}a|{\mathbf j} |}}{{4\omega \varepsilon }}\sin \theta \cos \theta {e^{ - jkr}}{e^{ - jl\phi }}\{ \frac{{j{k^2}}}{r}[{J_{l - 1}}(ka\sin \theta ) + {J_{l + 1}}(ka\sin \theta )]\\ + \frac{k}{{{r^2}}}[(2l + 1) \cdot {J_{l - 1}}(ka\sin \theta ) + (1 - 2l) \cdot {J_{l + 1}}(ka\sin \theta )]\} \end{array}$$

It can be seen that the z-component have the azimuth dependency of ${e^{ - jl\phi }}$, which means when l = ±3 and ±5, z-component generates OAM with topological charges of ±3 and ±5, respectively.

Figure 2(a) shows one practical configuration of 60 GHz OAM antenna. It consists of three layers. The top layer is a ring cavity with a circular slot. The middle layer is the transition layer with a threaded hole. The bottom layer is the feeding layer with two gradient waveguides. The metal ring cavity is fed by the gradient waveguide ports and the topological charge of generating OAM waves depends on excited waveguide ports. The main body of the OAM antenna is the air cavity as shown in Fig. 2(b). In order to enhance the performance of the OAM antenna, the optimized geometric dimensions are shown in Table 1. Full-wave numerical simulation (CST Microwave Studio) has been performed to validate the performances of this antenna. Due to the circular slot on the top wall of the air cavity, the electromagnetic field will radiate from the slot. The millimeter wave OAM antenna is symmetric along xoz plane. Here, only port 1 is excited. The simulated S parameters are showed in Fig. 2(c). Lower S parameters mean better system performances. One can clearly found that the return loss and isolation are both smaller than -10 dB from 58 to 62 GHz.

Table 1. Optimized dimensional parameters

View Table

Fig. 2. (a) Structure of the millimeter wave OAM antenna. (b) Structure of the main component air cavity. (c) S-parameters of the OAM antenna. |S₁₁| (red line) represents the return loss, and |S₂₁| (blue line) represents the isolation of the two waveguide ports.

Download Full Size | PDF

The near-field information including the phase and intensity distributions of the different electric components at 60 GHz are simulated and grouped in Figs. 3(a)-(f) for OAM mode l = -3. Here, the topological charges for x and y polarization are -2 and for the z polarization is -3, which are in good agreement with the mathematically calculated ones. The twisted phase distributions validate that the configurations in Fig. 3 can radiate the vortex beams. To illustrate the modal contents of the proposed millimeter wave OAM antenna, mode purities are calculated quantitatively through simulated data, and results are plotted in Figs. 3(g)-(i). It is clearly observed that the purities of the desired modes are all above 0.8, which are much higher than other disturbing modes.

Fig. 3. (a)-(c) Simulated phase distributions of the radiation electric field emitted from the millimeter wave OAM antenna for the x-, y- and z-polarized components, respectively. (d)-(f) Simulated x-, y- and z-polarized components intensities of the radiation electric field, respectively. (g)-(i) Mode purity of the emitted OAM waves x-, y- and z-polarized components, respectively.

Download Full Size | PDF

The proposed millimeter wave OAM antenna is fabricated and tested, whose parameters are the same with the simulated ones. The field distributions are measured by a near-field scan system as shown in Fig. 4(a), the probe is 70 mm (14$\mathrm{\lambda }$) in front of the OAM antenna aperture and scans with a scanning resolution of 2.4 mm in the transverse plane, and the overall scanning area is set to 60 mm × 60 mm. The OAM antenna consists of three layers as shown in Fig. 4(a). The measured reflection coefficient S₁₁ and isolation S₂₁ are smaller than -10 dB from 58 to 62 GHz as shown in Fig. 4(b). The measured results are in good agreement with the simulated ones as shown in Fig. 3(c). The measured x and y components at 60 GHz are depicted in Figs. 4(c),(d) and (f),(g), respectively. The magnitudes with a centered null and vortex phase distributions clearly illustrate that the OAM antenna can generate l = -2 mode for x and y polarization at 60 GHz. Slight discrepancies exist between the simulated results and measured ones due to the assembling errors imperfect fabrication, and measurement inaccuracy. To illustrate modal contents of the proposed OAM emitter, mode purity is calculated quantitatively through measured data, and results are plotted in Figs. 4(e) and (h). It is clearly observed that the predicted OAM mode l = -2 dominates the x and y components of the radiated electric field, and other OAM modes are negligible for the small weights. The simulated and experimental results agree well with each other and a null point is clearly observed in the main direction of wave propagation.

Fig. 4. (a) Experimental setup for near field measurement. (b) Measured S parameters of the fabricated OAM antenna. (c),(d) and (f),(g) are the measured phase and amplitude distributions for x and y polarization, respectively. (e) and (h) are the mode purities of the emitted OAM waves of x and y-polarization, respectively.

Download Full Size | PDF

The far-field radiation patterns of the OAM antenna are measured using a far-field system as shown in Fig. 5(a) to further verify the vortex-beam generation. One can see that the prototype is set as the transmitter and fixed on a turntable which can rotate 360° in horizontal plane, a standard horn antenna operates as the receiver. Figure 5 shows the simulated and measured 2D normalized radiation patterns of the l = -2 states for x-polarization in xoz plane and yoz plane, respectively. The simulated and experimental results coincide well with each other, and the radiation patterns with a null point in the normal direction, indicating that a phase singularity occurs at the center. For the antenna structure is completely symmetrical along the xoz plane, the currents generated from port 1 and port 2 are with the same magnitudes but opposite directions, so the distributions of the electric field generated by the two ports are also symmetrical along the xoz plane, which means the OAM with topological charge l = + 3 in z polarization and l = + 2 in x and y polarizations can be generated by feeding the port 2.

Fig. 5. (a) Experimental setup for far-field measurement. (b),(c) Simulated and measured normalized radiation patterns of l = - 2 mode for x polarization in xoz plane and yoz plane, respectively.

Download Full Size | PDF

In order to realize a much more attractive OAM mode multiplexing/demultiplexing system, a scheme of integrating multiple air cavities is proposed as shown in Fig. 6(a). Dual millimeter wave OAM antennas are used to generate modes of topological charges l = ±3 and l = ±5 in the z component and l = ±2 and l = ±4 in the x and y components at the operating frequency 60 GHz. Based on the OAM antenna with topological charges l = ±3 in z polarization above, we coaxially superimpose a larger ring cavity to generate l = ±5 modes OAM waves in z polarization and l = ±4 modes OAM waves in x and y polarizations. The optimized inner and outer radius are r_in = 9.13 mm, r_out = 11.9 mm, and the gap of the circular slot is gap = 0.7 mm. The simulated return loss S₁₁ and isolation loss S₂₁, S₃₁, S₄₁ are less than -10 dB at operating frequency 60 GHz as shown in Fig. 6(b). The phase and amplitude of the electric field distributions for x, y and z polarization are shown in Figs. 6(c)-(h), respectively. The magnitudes with centered null and vortex phase distributions clearly see that the OAM antenna can generate vortex beams at 60 GHz. The topological charges for x and y polarization are l = - 4 and for z polarization is l = - 5, which are in good agreement with the theoretical predictions. Fourier spectrum analysis of the measured field distributions are plotted in Figs. 6(i)-(k). Similar high OAM weights around 0.8 for the x, y and z components can be achieved. Besides, the OAM modes for l = + 5 in z component and l = + 4 in x and y components can be generated by feeding the port 2 with the same principle. The designer multiplexing millimeter wave OAM antenna is planar and the feed sources are gradient waveguide ports, which are able to integrate into the millimeter wave wireless communication systems.

Fig. 6. (a) OAM mode multiplexing antenna modal. (b) S-parameters of the OAM mode multiplexing antenna. (c)-(h) Simulated phase and amplitude distributions. (i)-(k) Mode purity of the emitted OAM waves.

Download Full Size | PDF

3. Discussion

In summary, we theoretically and experimentally demonstrated an easily realized OAM antenna based on the traveling-wave circular loop structure for efficiently multiplexing/demultiplexing multiple OAM modes at 60 GHz. The feeding networks are simply implemented using waveguide ports which are easy to integrate into millimeter wave communication systems. A prototype with OAM states l = ±3 carried by the z polarization and l = ±2 in the x and y polarizations at 60 GHz is fabricated and measured. Experimental nearfield distributions and far-field radiation patterns show excellent agreement with the simulated results. Furthermore, four coaxially propagating waves with OAM modes of l = ±3, ± 5 for z polarization component and l = ±2 and ±4 for x, y polarization components are investigated, respectively. Our work shows a promising perspective in millimeter wave wireless communication systems.

Funding

National Natural Science Foundation of China (62201500).

Disclosures

The authors declare no conflicts of interests.

Data availability

Data that support the findings of this study are available from the corresponding author upon reasonable request.

References

1. B. Li and H. Yin, “Peer-to-peer live video streaming on the internet: issues, existing approaches, and challenges [peer-to-peer multimedia streaming],” IEEE Commun. Mag. 45(6), 94–99 (2007). [CrossRef]

2. S. Misra, M. Reisslein, and G. Xue, “A survey of multimedia streaming in wireless sensor networks,” IEEE Commun. Surv. Tutorials 10(4), 18–39 (2008). [CrossRef]

3. L. F. de Souza Cardoso, F. C. M. Q. Mariano, and E. R. Zorzal, “A survey of industrial augmented reality,” Comput & Ind. Eng. 139, 106159 (2020). [CrossRef]

4. P. Chen, X. Liu, W. Cheng, and R. Huang, “A review of using Augmented Reality in Education from 2011 to 2016,” Innovations in Smart Learning13–18 (2017).

5. N. Cheng, F. Lyu, J. Chen, W. Xu, H. Zhou, S. Zhang, and X. Shen, “Big data driven vehicular networks,” IEEE Net. 32(6), 160–167 (2018). [CrossRef]

6. W. Shi, H. Zhou, J. Li, W. Xu, N. Zhang, and X. Shen, “Drone assisted vehicular networks: Architecture, challenges and opportunities,” IEEE Net. 32(3), 130–137 (2018). [CrossRef]

7. H. Ye, L. Liang, G. Y. Li, J. Kim, L. Lu, and M. Wu, “Machine learning for vehicular networks: Recent advances and application examples,” IEEE Veh. Technol. Mag. 13(2), 94–101 (2018). [CrossRef]

8. C. Jatoth, G. Gangadharan, U. Fiore, and R. Buyya, “SELCLOUD: a hybrid multi-criteria decision-making model for selection of cloud services,” Soft Computing 23(13), 4701–4715 (2019). [CrossRef]

9. S. K. Garg, S. Versteeg, and R. Buyya, “Smicloud: A framework for comparing and ranking cloud services,” in Fourth International Conference on Utility and Cloud Computing (IEEE, 2011), 210–218.

10. T. S. Rappaport, J. N. Murdock, and F. Gutierrez, “State of the art in 60-GHz integrated circuits and systems for wireless communications,” Proc. IEEE 99(8), 1390–1436 (2011). [CrossRef]

11. J. Reed, M. Vassiliou, and S. Shah, “The role of new technologies in solving the spectrum shortage [point of view],” Proc. IEEE 104(6), 1163–1168 (2016). [CrossRef]

12. Y. Yan, G. Xie, M. P. Lavery, H. Huang, N. Ahmed, C. Bao, Y. Ren, Y. Cao, L. Li, and Z. Zhao, “High-capacity millimetre-wave communications with orbital angular momentum multiplexing,” Nat. Commun. 5(1), 4876 (2014). [CrossRef]

13. L. Allen, M. W. Beijersbergen, R. Spreeuw, and J. Woerdman, “Orbital angular momentum of light and the transformation of Laguerre-Gaussian laser modes,” Phys. Rev. A 45(11), 8185–8189 (1992). [CrossRef]

14. V. V. Kotlyar, A. A. Almazov, S. N. Khonina, V. A. Soifer, H. Elfstrom, and J. Turunen, “Generation of phase singularity through diffracting a plane or Gaussian beam by a spiral phase plate,” J. Opt. Soc. Am. A 22(5), 849–861 (2005). [CrossRef]

15. E. Mari, F. Spinello, M. Oldoni, R. A. Ravanelli, F. Romanato, and G. Parisi, “Near-field experimental verification of separation of OAM channels,” Antennas Wirel. Propag. Lett. 14, 556–558 (2015). [CrossRef]

16. X. Meng, J. Wu, Z. Wu, L. Yang, L. Huang, X. Li, and T. Qu, “Design, fabrication, and measurement of an anisotropic holographic metasurface for generating vortex beams carrying orbital angular momentum,” Opt. Lett. 44(6), 1452–1455 (2019). [CrossRef]

17. Y. Zhang, Y. Lyu, H. Wang, X. Zhang, and X. Jin, “Transforming surface wave to propagating OAM vortex wave via flat dispersive metasurface in radio frequency,” Antennas Wirel. Propag. Lett. 17(1), 172–175 (2018). [CrossRef]

18. L. Jing, Z. Wang, Y. Yang, L. Shen, B. Zheng, F. Gao, H. Wang, E. Li, and H. Chen, “Diodelike spin-orbit interactions of light in chiral metasurfaces,” IEEE Trans. Antennas Propagat. 66(12), 7148–7155 (2018). [CrossRef]

19. Y. Yuan, K. Zhang, B. Ratni, Q. Song, X. Ding, Q. Wu, S. N. Burokur, and P. Genevet, “Independent phase modulation for quadruplex polarization channels enabled by chirality-assisted geometric-phase metasurfaces,” Nat. Commun. 11(1), 4186 (2020). [CrossRef]

20. Y. Wang, Y. Yuan, G. Yang, X. Ding, Q. Wu, Y. Jiang, S. N. Burokur, and K. Zhang, “Perfect control of diffraction patterns with phase-gradient metasurfaces,” ACS Appl. Mater. Interfaces 14(14), 16856–16865 (2022). [CrossRef]

21. X. Gao, S. Huang, Y. Song, S. Li, Y. Wei, J. Zhou, X. Zheng, H. Zhang, and W. Gu, “Generating the orbital angular momentum of radio frequency signals using optical-true-time-delay unit based on optical spectrum processor,” Opt. Lett. 39(9), 2652–2655 (2014). [CrossRef]

22. Y. Pan, S. Zheng, J. Zheng, Y. Li, X. Jin, H. Chi, and X. Zhang, “Generation of orbital angular momentum radio waves based on dielectric resonator antenna,” Antennas Wirel. Propag. Lett. 16, 385–388 (2017). [CrossRef]

23. L. Kang, H. Li, J. Zhou, S. Zheng, and S. Gao, “A mode-reconfigurable orbital angular momentum antenna with simplified feeding scheme,” IEEE Trans. Antennas Propagat. 67(7), 4866–4871 (2019). [CrossRef]

24. Q. Liu, Z. N. Chen, Y. Liu, F. Li, Y. Chen, and Z. Mo, “Circular polarization and mode reconfigurable wideband orbital angular momentum patch array antenna,” IEEE Trans. Antennas Propagat. 66(4), 1796–1804 (2018). [CrossRef]

25. G.-B. Wu, K. F. Chan, S.-W. Qu, K. F. Tong, and C. H. Chan, “Orbital angular momentum (OAM) mode-reconfigurable discrete dielectric lens operating at 300 GHz,” IEEE Trans. THz Sci. Technol. 10(5), 480–489 (2020). [CrossRef]

26. E. Berglind and G. Björk, “Humblet's decomposition of the electromagnetic angular moment in metallic waveguides,” IEEE Trans. Microwave Theory Techn. 62(4), 779–788 (2014). [CrossRef]

27. S. Zheng, W. Zhang, Z. Zhang, X. Jin, H. Chi, and X. Zhang, “Generation and propagation characteristics of electromagnetic vortices in radio frequency,” Photonics Res. 4(5), B9–B13 (2016). [CrossRef]

28. Z. Zhang, S. Xiao, Y. Li, and B.-Z. Wang, “A circularly polarized multimode patch antenna for the generation of multiple orbital angular momentum modes,” Antennas Wirel. Propag. Lett. 16, 521–524 (2017). [CrossRef]

29. Y. Zhang, Q. Zhang, C. H. Chan, E. Li, J. Jin, and H. Wang, “Emission of orbital angular momentum based on spoof localized surface plasmons,” Opt. Lett. 44(23), 5735–5738 (2019). [CrossRef]

30. Y. Chen, S. Zheng, H. Chi, X. Jin, and X. Zhang, “Half-mode substrate integrated waveguide antenna for generating multiple orbital angular momentum modes,” Electron. Lett. 52(9), 684–686 (2016). [CrossRef]

31. J. Ren, K. W. Leung, D. Q. Liu, K. M. Luk, and J.-F. Mao, “Orbital angular momentum radiator multiplexing electromagnetic waves in free space,” Opt. Express 28(1), 345–359 (2020). [CrossRef]

32. N. Yu, P. Genevet, M. A. Kats, F. Aieta, J.-P. Tetienne, F. Capasso, and Z. Gaburro, “Light propagation with phase discontinuities: generalized laws of reflection and refraction,” Science 334(6054), 333–337 (2011). [CrossRef]

33. N. Yu and F. Capasso, “Flat optics with designer metasurfaces,” Nat. Mater. 13(2), 139–150 (2014). [CrossRef]

34. G.-B. Wu, K. F. Chan, K. M. Shum, and C. H. Chan, “Millimeter-wave holographic flat lens antenna for orbital angular momentum multiplexing,” IEEE Trans. Antennas Propagat. 69(8), 4289–4303 (2021). [CrossRef]

35. Z. He, Y. Wang, X. Wang, A. S. Kupriianov, V. R. Tuz, and V. I. Fesenko, “Multi-band orbital angular momentum mode-division multiplexing by a compact set of microstrip ring-shaped resonator antenna,” Opt. Express 30(26), 46209–46226 (2022). [CrossRef]

36. Z.-G. Guo and G.-M. Yang, “Radial uniform circular antenna array for dual-mode OAM communication,” Antennas Wirel. Propag. Lett. 16, 404–407 (2017). [CrossRef]

37. P.-Y. Feng, S.-W. Qu, and S. Yang, “OAM-generating transmitarray antenna with circular phased array antenna feed,” IEEE Trans. Antennas Propagat. 68(6), 4540–4548 (2020). [CrossRef]

38. D. Wu, Z. Zhang, G. Fu, X. Shi, L. Yang, and X. Li, “Rotman lens-fed antenna for generating multiple orbital angular momentum (OAM) modes with gain enhancement,” IEEE Access 8, 29891–29900 (2020). [CrossRef]

Parameter	value	Parameter	value	Parameter	value	Parameter	value
l₁	11.3 mm	r₃	2.16 mm	d₃	3.7 mm	w_s	0.63 mm
w₁	2.8 mm	a	3.76 mm	h₁	1.18 mm	w₂	1.3 mm
l₂	13.5 mm	b	1.88 mm	h₂	1.5 mm	w₃	0.78 mm
r₁	4.92 mm	d₁	2.2 mm	h₃	4.38 mm	l_g	2.7 mm
r₂	7.72 mm	d₂	1.9 mm	h₄	0.68 mm	w_g	0.6 mm

Generation of orbital angular momentum multiplexing millimeter waves based on a circular traveling wave antenna

Abstract

1. Introduction

2. Results

3. Discussion

Funding

Disclosures

Data availability

References

Data availability

Cited By

Figures (6)

Tables (1)

Equations (9)

Optics Express