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Abstract
A metasurface (MS) is highly useful for improving the performance of patch antennae and reducing their size due to their inherent and unique electromagnetic properties. In this paper, a compact and broadband antenna based on a step-shaped metasurface (SMS) at an operating frequency of 4.3 GHz is presented, which is fed by a planar monopole and enabled by selecting an SMS with high selectivity. The SMS consists of an array of metallic step-shaped unit cells underneath the monopole, which provide footprint miniaturization and bandwidth expansion. Numerical results show that the SMS-based antenna with a maximum size of (where λ0 is the operating wavelength in free space) exhibits a 22.3% impedance bandwidth (S11 < -10 dB) and a high gain of more than 7.15 dBi within the passband. Experimental results at microwave frequencies verify the performance of the proposed antenna, demonstrating substantial consistency with the simulation results. The compact and broadband antenna therefore predicts numerous potential applications within modern wireless communication systems.
© 2017 Optical Society of America
1. Introduction
Metasurfaces (MSs), which are the two-dimensional equivalents of metamaterials, have become a research hotspot over the past decade due to their unusual electromagnetic responses that are not observed within natural materials [1–3]. Being a sub-field of metamaterials [4], MSs are often composed of periodically arranged metallic unit cells characterized by a homogeneous or inhomogeneous subwavelength, and have been widely employed within various applications due to their inherent advantages regarding an enhanced performance and compact planar structure with a low profile [5–7]. MSs consist of a few different types, including electromagnetic band-gap structures (EBGs) [8,9], high-impedance surfaces (HISs) [10–12] or reactive impedance surface (RISs) [13], and artificial magnetic conductors (AMCs) [14–17].
It has been observed that MSs have the ability to effectively improve the functional performance of antennae in many aspects, including circular polarization [18], a low-profile [19], radiation pattern reconfiguration [20], and directivity enhancement [21]. Recently, substantial research efforts have been devoted to miniaturize antenna dimensions with the assistance of MSs. Previous studies have indicated that the fabrication of a prototype patch antenna on an RIS substrate can achieve a miniaturization of λ0/10 [22]. The RIS consists of a periodic array of square patches printed upon a metal-backed dielectric material. To further investigate the properties of RISs, different types of patch antennae on RIS substrates have been studied to examine antenna size miniaturization and improve their circularly polarized (CP) radiation [23,24]. Furthermore, Xu and Cai et al. [25,26] demonstrated that the combination of an MS with a meta-resonator can reduce the antenna size. Both of these materials can provide a low-resonance frequency in this way enabling antenna miniaturization. Dong et al. [27] introduced miniaturized patch antennas loaded with CSRRs over an RIS. A smaller antenna is subsequently obtained when microstrip-line-slot fed or patch antennae are loaded with composite right/left-handed (CRLH) mushroom-like structures [28,29]. A layered MS is often utilized to miniaturize the size of a patch antenna [30]. Simulations and experimental results demonstrate that the size of a patch antenna can be reduced by 67% when loaded by an MS. Wearable antennae play an important role in medical devices. For example, Raad et al. [31] presented a flexible and compact antenna composed of a monopole antenna integrated with a compact AMC ground plane for wearable telemedicine applications. Subsequently, an improved wearable antenna with a compact footprint was proposed by Jiang et al. [32], which was based on an I-shaped anisotropic MS for operation within a medical body-area network band. More recently, a class of compact and wideband MS-based antennas was designed using various interdigitated capacitive (IC) MSs [33], and it is shown that the polarization characteristics of the antenna can be adjusted by these IC MSs.
In this paper, we propose and experimentally fabricated a miniaturized footprint antenna with bandwidth enhancement characteristics based on MSs. The antenna is composed of a microstrip-fed monopole antenna and an MS with an array of unit cells, and the metallic sheet backing is an MS with a step-shaped slot. Simulation results demonstrate that a step-shaped metasurface (SMS) attached to the back of a monopole antenna can decrease the resonant frequency and improve the performance of the antenna compared with a monopole. The organization of the paper is as follows. Firstly, the structure of the SMS is presented and the characteristics of the unit cell are analyzed in Section 2. Then, the SMS assisted antenna is designed and simulation results of its performance are given in Section 3. Afterwards, in Section 4, the formulae for a transmission-line model are used to analyze the resonance characteristics of the SMS unit cell and a comparison with the numerical results of the dispersion relationship is discussed in Section 4. A prototype antenna is fabricated and experimental results are given in Section 5. Finally, conclusions of the study are briefly provided in Section 6.
2. Theoretical and SMS unit cell analysis
Numerical simulations of the unit cell have been conducted using Ansys’ HFSS software to analyze the electromagnetic response of an infinite SMS. Figure 1(a) illustrates the simulation model of the unit cell, which is illuminated by a transverse electric and magnetic (TEM) planar wave incident along the negative z-direction, while perfect electric conductor (PEC) and perfect magnetic conductor (PMC) boundaries are applied along the x- and y-directions, respectively. The effective equivalent circuit model for the SMS unit cell is depicted in Fig. 1(b), and its resonant frequency can be computed by [31] as follows:
where Ls and Cs are the equivalent inductance and capacitance of the step-shaped metallic patch printed onto the substrate. Under this circumstance, Ld represents the inductance of the substrate, the value of which is primarily dependent on the permittivity and thickness of the substrate. The structure can be miniaturized by increasing the quantity of the denominator of Eq. (1) to decrease the resonant frequency. Obviously, increasing Ld will lead to a higher profile, which is undesirable. The bandwidth of the unit cell can be expressed by Eq. (2) [31]:
Fig. 1 (a) Configuration of an SMS unit cell. (b) Effective equivalent circuit model illuminated by a plane wave incident along the negative z-direction.
It is evident from Eq. (2) that the bandwidth will be narrower following an increase of Ls rather than Cs. Hence, we focus on increasing the value of the shunt capacitance Cs to achieve a more miniaturized structure in this article.
To illustrate the advantage of the SMS unit cell, we have also simulated the reflection phase of a conventional unit cell, which has the same geometric parameters with the proposed structure but without steps. We can clearly observe in Fig. 2(a) that the SMS unit has a lower resonant frequency at zero reflection phase. The dependence of the reflection phase on the geometric parameters are additionally simulated and are plotted in Figs. 2(b)–2(d). It is therein apparent that the resonant frequency fr moves to a lower frequency when the gap width (s) is reduced, or when the step length (fh) or step number (steps) is increased. However, the fractional bandwidth which is defined by Eq. (3) will be consequently narrower.
Here, f2, f1 and f0 are the frequencies when the reflection phase is ±90° and 0°. For example, the fractional bandwidths of 13.8% and 10.3% corresponds to the unit cell with 1 and 5 steps, respectively, as shown in Fig. 2(d). This conclusion can also be verified by Eq. (2), and the results from the above discussion can provide useful guidance for practical design. In the following simulation, we set the parameters as gap width s = 0.2 mm, step length fh =1 mm and 3 steps are adopted as a compromise between a lower resonant frequency and bandwidth.Fig. 2 Reflection phase with (a) a conventional unit cell and SMS unit cell, (b) a different gap width (s), (c) a different step length (fh), and (d) different step numbers.
3. Antenna design and performance analysis
The structure of the proposed antenna is shown in Fig. 3. It is composed of a planar monopole (as the source antenna) and an SMS with a metallic sheet backing. The spacing between the two structures is a foam substance with a thickness of 0.5 mm. The monopole antenna consists of a hexagon patch fed by the 50-Ω microstrip line. The microstrip patch and the partial ground plane are printed onto the top and bottom faces of a dielectric substrate (Rogers 3003, or RO3003) whose relative dielectric constant is εr = 3 and lost tangent is tan δ = 0.0013. The SMS consists of 3×6 unit cells, wherein each unit cell is formed by a square patch with a step-shaped slot that is printed onto the RO3003 substrate. The overall dimension of the antenna is 0.42 with a low profile of 0.065λ0 at an operating frequency of 4.3 GHz (λ0= 69.8 mm).
Fig. 3 Structural illustration of the SMS antenna. The optimized dimensions are Ax = 25, Ay = 20, Lx = 62.4, Ly = 33, g = 0.4, al = 12.2, aw = 3.6, gl = 10, r = 5.5, d = 0.5, ha = 1.6, hs = 2.5, all in millimeters.
As a comparison, reflection coefficients of the monopole and the SMS antenna are simulated and are illustrated in Fig. 4(a). It is observed that the resonant frequency can be reduced from 4.86 GHz to 4.075 GHz and 4.685 GHz when the monopole is loaded with an SMS. In addition, an agreeable impedance match is achieved and the bandwidth of S11 < -10dB is increased by 22.3%, that is, from 3.82 GHz to 4.78 GHz with a central frequency of 4.3 GHz. The influence of the gap width(s) and step length(fh) on S11 are investigated, the results are plotted in Figs. 4(b) and 4(c), respectively. It is observed that the increase of gap width and step length cause a redshift of the resonant frequency, which agrees well with the results obtained from the eigenmode analysis of a unit cell in Fig. 2. Figure 4(d) displays the simulated gain for both the monopole and the SMS antenna, and it is clear that the antenna gain can be significantly improved when the SMS structure is positioned at a distance d below the monopole. The average gain has been enhanced from 1.9 dBi to 7.7 dBi within the frequency band where S11 < -10 dB. It should be noted that the matching of impedance for the SMS antenna can also be tuned by adjusting the distance d between the monopole and the SMS structure, which will impact the coupling degree of electromagnetic energies.
Fig. 4 Simulations of S11 with (a) a monopole and SMS antenna, (b) a different gap width s, and (c) a different step length. (d) Simulated gain for a monopole and an SMS antenna.
Figures 5(a) and 5(b) depict the simulated E- and H-plane radiation patterns for the SMS antenna at 4.3 GHz. For comparison purposes, the radiation patterns of the monopole alone are also shown in the same figure. Relative to the bi-directional radiation pattern of the monopole alone, the energy of the SMS antenna is more preferentially radiated broadside to the antenna, and the radiation patterns become strongly directional towards the positive z-direction. The simulated half-power beam widths (HPBWs) of the proposed antenna are 88° and 64° in the E-plane and H-plane at 4.3 GHz, respectively. Moreover, the FB (front-to-back) ratio increases to approximately 12 dB in the frequency band of interest, which is remarkably higher than that of the monopole, which exhibits a gain of only 0.6 dB in the same frequency band as shown in Fig. 5(c). From Fig. 5(c), we can also observe that the AR < -3 dB possess a dual-band including bandwidths of 220 MHz and 100 MHz, ranging from 3.85 to 4.07 GHz and 4.73 to 4.83 GHz.
Fig. 5 A comparison between the monopole and the SMS antenna. (a) E-plane radiation patterns at 4.3 GHz. (b) H-plane radiation patterns at 4.3 GHz. (c) FB ratio and the axial ratio.
4. Radiation mechanism of the SMS antenna
The radiating mechanism of the proposed SMS-enabled antenna is illustrated in this section. As observed in Fig. 4(a), the SMS antenna has two resonant modes in the passband, and the E-field distributions of these resonant modes at 4 and 4.6 GHz are simulated, as depicted in Figs. 6(a) and 6(b). As a comparison, E-field distribution of TM10 and TM20 modes for a conventional patch antenna predicted by the classical cavity model, is also shown in the same figures indicated by grey arrows. We clearly observe that E-field distribution of the SMS antenna at 4 and 4.6 GHz are similar to those of the TM10 and TM20 modes of a conventional rectangular microstrip antenna, except for the radiation from the slots of the SMS structure. The simulated E-field distribution of the TM10 and TM20 modes for the monopole with full grounding in y = 0 plane appear at 4 and 8 GHz (not shown here). This indicates that, with the assistance of the SMS, the second mode moves to a lower frequency, which give rise to a miniaturization of the antenna.
Fig. 6 Simulated E-field distribution of the SMS antenna and expected E-field distribution based on the cavity model for (a) TM10 and (b) TM20 modes in the x = 33 mm plane at 4 and 4.6 GHz, respectively.
An extra resonance for the antenna system is yielded when utilizing the SMS structure. This phenomenon reveals that the surface waves propagating along the top layer of the SMS are excited, and these results can be used to improve antenna performance. When the height of the SMS substrate hs is much smaller than the wavelength λ0 in a vacuum, the two resonant frequencies for the TM10 and TM20 modes can be analyzed with regard to transmission-line model theory. To reveal the nature of the surface wave resonance, the dispersions for the x- and y-directions are calculated using the eigenmode solver of HFSS, the results of which are shown in Fig. 7. As is illustrated in Fig. 7, the SMS unit cell is enclosed by an air box, periodic boundary conditions are applied to surround the air box, and a perfectly matched layer (PML) boundary is applied atop the air as an absorbent material. The parameters Px and Py indicate the period of the unit cell in the x- and y-directions. There is no fringing field at the open edges of the SMS array along the x-direction, and therefore the extension length ΔL is zero. The surface wave resonance for the TM10 mode can be described by the following Eq. (4) [29]:
where βm represents the propagation constant and Nx represents the number of unit cells in the x-direction (Nx = 3 in this paper). The calculated resonant frequency for the TM10 mode is 3.89 GHz (βmPx/2π = 1/6), which can be achieved from the dispersion relationship shown in Fig. 7 (i.e., the blue line). Owing to the existence of the fringing field at the open edges of the array in the y-direction, the extension length ΔL can be expressed aswhere W = NyPy is the effective width in the y-direction (Ny = 6 is the number of unit cells in the y-direction), hs is the height of the SMS substrate, and the effective dielectric constant εreff is given aswhere k0 is the wave vector of the vacuum, f is the antenna working frequency and c is the velocity of light. The resonant frequency for the TM20 mode is determined byTherefore, the resonant frequency for the TM20 mode is 4.63 GHz, which can be predicted from the dispersion curve in Fig. 7 when βmPy/2π = 0.145 (i.e., the red line). The two surface wave resonances for the TM10 and TM20 modes, which are extracted from the dispersion curves, are similar to the first and second resonant frequencies (4.075 and 4.685 GHz) of the SMS antenna, respectively. In summary, the radiation mechanism of the SMS antenna can be satisfactorily explained using a transmission-line model.
To understand the mechanism through which the broadside radiation is generated by the proposed antenna, we have plotted the surface current distribution along the top surface of the SMS at 4.6 GHz is shown in Fig. 8(a). It can be observed that the dominant surface currents are in-phase and flow along the negative x-direction. In contrast, the surface currents of the monopole alone are out-of-phase. These observations demonstrate that the TM20 mode can be excited to obtain the radiation within the broadside direction. The broadside radiation of the TM20 mode is further explained by the radiation that is emitted from the step-shaped slots of the SMS. Fig. 8(b) displays the simulated distribution of the electric fields of the SMS on the opposite side at 4.6 GHz. The strong electric fields are mainly concentrated in the vicinity of the twist slots. The radiation corresponding to the existence of the radiating slots can be attributed to the magnetic current induced by the electric fields, which generates broadside radiation.
Fig. 8 (a) Simulated surface current distribution and (b) distribution of electric fields of the SMS antenna at 4.6 GHz.
5. Experimental results
To verify the performance of the proposed SMS-based antenna, an antenna prototype has been fabricated as shown in Fig. 9. Due to the unavailability of a sufficiently thick RO3003 material, we used F4B (εr = 2.65) as the substrate. To facilitate the fabrication and experiment, ax, al and gl are set to 30, 17.2 and 15 mm, respectively, while the other parameters are equivalent to those in Fig. 3. As illustrated in Fig. 10(a), the fabricated antenna prototype is composed of two components, i.e., the monopole (front) and MS (back), which are insulated by a thin foam. The device is connected to a Vector Network Analyser (PNA-L, N5234A) via coaxial cables to measure the reflection coefficient, gain and radiation patterns through a far-field measurement system situated within an anechoic chamber.
Fig. 9 Photographs of (a) the fabricated copper ground plane backed step-shaped MS, as well as (b) the top view and (c) the bottom view of the monopole antenna.
Fig. 10 (a) Experimental setup. (b) The simulated and measured reflection coefficients for S11, as well as the (c) E-plane and (d) H-plane radiation patterns at 4.4 GHz.
Figure 10(b) displays the simulated and measured reflection coefficient and gain. Overall, the measured impedance bandwidth for the -10 dB reflection coefficient spans from 3.86 to 4.92 GHz or 24%, and the measured impedance bandwidth spans from 3.87 to 4.85 GHz or 23%. It is apparent that the measured and simulated results are in close agreement, except that the second resonance dip position is slightly shifted towards a higher frequency. The simulated and measured average gain in the targeted frequency band is roughly 7.38 and 7.07 dBi, respectively. A maximum gain of 7.26 dBi is measured at 4.2 GHz over the impedance bandwidth. The simulated and measured radiation patterns of the antenna at a frequency of 4.4 GHz in the E- and H-planes are presented in Figs. 10(c) and 10(d), respectively, for which a reasonable agreement is observed between the measured and simulated patterns. The simulated and measured E-plane radiation patterns have HPBWs equal to 82° and 86°. Meanwhile, in the H-plane, the HPBWs of the simulated and measured radiation patterns are 53° and 58°, respectively. It should be noted that a minor disagreement between the simulated and measured patterns is observed at a measurement point of 330°, which is most likely due to error within the fabrication tolerance and measurement system, as well as in the foam between the two components.
6. Conclusion
In summary, a compact and broadband antenna enabled by an SMS has been proposed. Compared with a monopole alone, the antenna has lower frequencies when S11 < -10 dB, and the miniaturized total structure is 0.42 with a low profile of 0.065λ0. Moreover, the antenna achieves an impedance bandwidth of 22.3% from 3.82 to 4.78 GHz with an average gain of 7.7 dBi, and it exhibits the desired broadside radiation. Through an analysis of the reflection phase of the SMS unit cell, the resonant characteristics can be flexibly adjusted by tuning the dimensions of the slots, thereby providing a guideline for the actual design. An antenna prototype has been fabricated. The reflection coefficient, gain and radiation patterns agree with those of the simulations. The proposed compact antenna would be a good candidate for potential application within wireless communication systems.
Funding
National Natural Science Foundation of China (Grant Nos. 61261002, 61461052, 11564044); Specialized Research Fund for the Doctoral Program of Higher Education (Grant Nos. 20135301110003, 20125301120009); the Key Program of Natural Science of Yunnan Province (Grant Nos. 2013FA006, 2015FA015); Spectrum Sensing and borderlands Security Key Laboratory of Universities in Yunnan (C6165903); Yunnan University's Research Innovation Fund for Graduate Students.
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