Real-time normalization and nonlinearity evaluation methods of the PGC-arctan demodulation in an EOM-based sinusoidal phase modulating interferometer

Shihua Zhang; Benyong Chen; Liping Yan; Zheyi Xu

doi:10.1364/OE.26.000605

1. Introduction

The phase modulation of an electro-optic modulator (EOM) is produced by a voltage-induced change in the refractive index of the electro-optic material (e.g. lithium niobate [1], silica [2] and polymers [3]). As having the properties of high modulating frequency and high intrinsic modulation bandwidth, the EOM is widely used in the phase modulating interferometers. L. Yan et al. applied a period sawtooth voltage to an EOM to construct a linear phase modulated dual-homodyne interferometer without polarizing optics, and two sinusoidal interference signals were obtained for zero-crossing phase detecting [4]. This interferometer can eliminate the periodic nonlinear error and has the ability of anti-interference, while the sawtooth signal is not appropriate to high frequency modulation, thus the measurement speed is limited. An alternative to linear phase modulation is sinusoidal phase modulation, which facilitates high-speed measuring system. For instance, U. Minoni et al. proposed a high-frequency sinusoidal phase modulating interferometer (SPMI) with an EOM operating at frequencies up to 10 MHz, and the target can speed up to 1 m/s [5].

For conventional SPMI, the sinusoidal phase modulation can be realized by adding a piezoelectric (PZT) modulator on one arm of the interferometer [6–8], or applying a current modulation to a laser diode [9]. Compared with phase modulation with EOM, the PZT modulation cannot realize high speed measurement because of the limited vibrating frequency of the PZT, and the laser diodes current modulation is accompanied with optical intensity modulation which will cause instability and distortion of phase demodulation output. Coupled with the merits of stable performance and large dynamic range, sinusoidal phase modulation with EOM has been used in optical feedback interferometer [10] and self-mixing interferometer [11] for high speed displacement measurement with large range.

With sinusoidal phase modulation, a phase carrier is generated which up-converts the desired phase signal onto the sidebands of the carrier frequency and the corresponding homodyne detection scheme is phase generated carrier (PGC) demodulation. Due to the distinct advantages of simple construction, high sensitivity and the capability of real-time demodulation, the PGC demodulation has been used to fiber optical interferometric sensors [7, 12] and the measurements of displacement, vibration and surface topography [13–15]. Generally, the fundamental and second harmonic of the interference signals are chosen to acquire a pair of quadrature components containing the desired phase signal. The PGC algorithms employing the differential-and-cross-multiplying approach (PGC-DCM) [16] and the arctangent approach (PGC-Arctan) [17] have been well developed to recover the phase from the quadrature components. The PGC-DCM approach has relatively low harmonic distortion, while its result is influenced by the light intensity disturbance (LID) and the visibility of the interference fringes. The scheme was improved with the PGC-Arctan approach, which is immune to the changes of the LID and the visibility of the interference fringes [18]. However, a phase delay between the carrier component of the detected interference signal and the carrier signal, a deviation of the phase modulation depth used for signal processing from the actual value and the non-ideal performance of the low pass filters will introduce periodic nonlinearity in phase demodulation.

Several methods have been proposed to eliminate the influences of the phase delay and the phase modulation depth. K. Wang et al. employed the orthogonal detection method to obtain the phase delay introduced by the fiber-optic interferometric sensor (FIOS), while the signals with non-perfect orthogonality will introduce an error for phase detection [19]. A phase compensator based on PGC-DCM approach was designed by S. C. Huang et al. to compensate the phase delay caused by the transmission of the laser light in the fibers for FOISs, while the phase delay correction was carried out offline and the compensate angle cannot be changed automatically when the phase delay varied [20]. In our previous work, a real-time phase delay compensator was proposed to track the change of the phase delay and the proposed compensator is appropriate to both PGC-DCM approach and PGC-Arctan approach [21]. Nevertheless, to ensure the PGC demodulating work properly, the methods mentioned above required the phase modulation depth at certain value (2.37 rad for PGC-DCM approach and 2.63 rad for PGC-Actan approach [22]) or the modulation depth used for signal processing equal to the actual value strictly. J. He et al. presented an ameliorated approach based on arctangent function and differential-self-multiplying (DSM) to remove the effect of the phase modulation depth [23], while the results of differential operation will be impacted seriously by the noise of the interference signal in ultra-low speed or static measurement. A. V. Volkov et al. proposed a phase modulation depth evaluation and correction technique based on an integral control feedback to provide the phase modulation depth stabilization to a certain predefined optimal value [24], while except for the phase modulation depth, the carrier phase delay and some other factors may occur simultaneously in the experiments. Normalizing the quadrature components obtained by low pass filters can eliminate the influences induced by the factors mentioned above at a time. C. Ni et al. presented an ellipse fitting method which can estimate the parameters to correct the measurement error by moving the target to produce several periods of interference signals [25]. However, when the target is in stationary or the movement of the target is much less than half of a laser wavelength, the ellipse fitting will not be implemented correctly or the result of the ellipse fitting is inaccuracy. Considering that the phase delay varies with the optical path, and the phase modulation depth is also not a constant during the measurement (e.g. the phase modulation depth of the EOM will drift with the half-wave voltage which is related to the environmental temperature), a universal method which can depress the influences mentioned above simultaneously and timely is on the urgent need.

In this work, a real-time normalization method of PGC-Arctan demodulation is proposed based on a modified SPMI with an EOM to suppress the periodic nonlinearity. A fixed-phase-difference detection method for evaluating the nonlinearity of the phase demodulation in real time is also presented. The theory of real-time normalization and evaluation of the periodic nonlinearity in PGC-Arctan demodulation is described in section 2. In section 3, the verifying experiments and results are given.

2. Theory

2.1 PGC-Arctan demodulation with real-time normalization

With sinusoidal phase modulation, the optical path difference of the interferometer is modulated and a high-frequency sinusoidal phase carrier is introduced to the interferometer. Supposing the carrier signal is

V_{ω_{c}} (t) = A \cos ω_{c} t

where A and ω_c are the amplitude and frequency of the signal. The interference signal is given by

S (t) = S_{0} + S_{1} \cos [z \cos (ω_{c} t - θ) + φ (t)]

where S₀ and S₁ are the amplitudes of the DC component and the AC component, respectively; z is the phase modulation depth, θ is the phase delay of the carrier component and φ(t) is the phase to be demodulated. Expending Eq. (1), we can obtain

\begin{array}{l} S (t) = S_{0} + S_{1} \cos φ (t) [J_{0} (z) + 2 \sum_{m = 1}^{\infty} {(- 1)}^{m} J_{2 m} (z) \cos 2 m (ω_{c} t - θ)] \\ + S_{1} \sin φ (t) [- 2 \sum_{m = 1}^{\infty} {(- 1)}^{m} J_{2 m - 1} (z) \cos (2 m - 1) (ω_{c} t - θ)], \end{array}

where J₍₂_m_-1)(z) and J₂_m(z) denote the odd- and even-order Bessel functions, respectively. The schematic of PGC-Arctan demodulation is shown in Fig. 1. Firstly, multiplying the interference signal S(t) with the carrier Vω_c(t) and the second-harmonic carrier V2ω_c(t), respectively. Then, filtering out the high-frequency signals including the fundamental carrier and all harmonic carrier frequencies by low-pass filters (LPFs) and dividing the theoretical value of the Bessel functions, a pair of quadrature components can be obtained as

P_{1} (t) = \frac{L P F [S (t) \cdot V_{ω_{c}} (t)]}{J_{1} (z^{'})} = - \frac{J_{1} (z)}{J_{1} (z^{'})} K_{1} S_{1} A \cos θ \sin φ (t)

P_{2} (t) = \frac{L P F [S (t) \cdot V_{2 ω_{c}} (t)]}{J_{2} (z^{'})} = - \frac{J_{2} (z)}{J_{2} (z^{'})} K_{2} S_{1} A \cos 2 θ \cos φ (t)

where K_i (i = 1,2) is the total gain of the multiplier and the low-pass filter, z′ is the theoretical value of the phase demodulation depth used for signal processing. Obviously, the Lissajous figure of P₁(t) and P₂(t) should be a circle if K₁ = K₂, z = z′ and θ = kπ (k = 0,1,2,…). However, in experiment, K₁ is not exactly equal to K₂ due to the non-ideal performance of the multipliers and the LPFs. Moreover, a deviation of the phase demodulation depth z′ used for signal processing from the actual value z and a phase delay between the carrier and the detected interference signal often occurs, and they may vary with the environmental temperature and the measuring displacement. All of these factors make the Lissajous figure of P₁(t) and P₂(t) to be an ellipse, and a periodic nonlinearity will be induced to the demodulated phase. Supposing that the major axis and minor axis of the ellipse are α₁ = −J₁(z)K₁S₁Acosθ/J₁(z′) and α₂ = −J₂(z)K₂S₁Acos2θ/J₂(z′), respectively, they are defined as the normalized parameters of the quadrature components. If α₁ and α₂ are estimated by ellipse fitting of the Lissajous figure or calculating the amplitudes of P₁(t) and P₂(t), Eqs. (4) and (5) can be normalized as

Q_{1} (t) = \frac{P_{1} (t)}{α_{1}} = \sin φ (t)

Q_{2} (t) = \frac{P_{2} (t)}{α_{2}} = \cos φ (t)

Therefore, with real-time normalization, all the factors affecting the phase demodulation are eliminated and the Lissajous figure of the quadrature components transforms to a unit circle. After the operations of division and arctangent, the demodulated phase is described as

φ (t) = arc \tan \frac{Q_{1} (t)}{Q_{2} (t)}

It can be deduced that, the method of real-time normalization can also eliminate the influences induced by light intensity disturbance (LID) and the visibility varying of interference fringes in the DCM approach of PGC demodulation.

Fig. 1 Schematic of PGC-Arctan demodulation with real-time normalization.

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In order to realize real-time normalization in PGC demodulation, a modified EOM-based SPMI consisting of a monitor interferometer and a probe interferometer is constructed. The two interferometers share a reference corner cube which is mounted on a stage moving back and forth with an ultra-low speed. Then periodic interference signals are generated for both the monitor and probe interferometers, and the normalized parameters can be always estimated to realize real-time normalization in PGC demodulation whether the target is in dynamic or static state.

The schematic of the modified SPMI is shown in Fig. 2. The laser beam emitted from a He-Ne laser is linearly polarized by a polarizer (P) along the vertical direction which is the electro-optic crystal's extraordinary axis direction, and split into two parts by a beam splitter (BS₁). The reflected part is introduced to the common reference arm consisting of an EOM and a reference corner cube (CC₁) mounted on a slowly moving stage, then propagates to another beam splitter (BS₂). The transmitted part passes through the third beam splitter (BS₃) and propagates to the measurement corner cube (CC₂). The reflected beams from BS₂ and BS₃ are combined at the forth beam splitter (BS₄) and interfere with each other, then the interference signal of the monitor interferometer is generated and detected by a photodetector (PD₁). The transmitted beams from BS₂ and CC₂ are combined at BS₁, and the interference signal of the probe interferometer is generated and detected by two photodetectors (PD₂, PD₃) which are placed at an interval of a quarter of interference fringe.

Fig. 2 Schematic of the modified EOM-based SPMI.

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If the modulating signal applied to the EOM is

V (t) = β V_{ω_{c}} (t) = β A \cos ω_{c} t

where β is the amplification factor of a high voltage amplifier (HVA) used to drive the EOM. Then the phase shift caused by the EOM is given by

φ_{E O M} = \frac{π β A}{V_{π}} \cos ω_{c} t

where V_π is the half-wave voltage (the voltage required to shift the output phase by π radians) of the EOM. Assuming that the initial optical path differences of the monitor interferometer and the probe interferometer are l₁ and l₂, respectively, the detected interference signals of the monitor and probe interferometers can be expressed as

\begin{array}{l} S_{1} (t) = S_{01} + S_{11} \cos {\frac{π β A}{V_{π}} \cos (ω_{c} t - θ_{1}) + \frac{2 π}{λ} [l_{1} + 2 l (t)]} \\ = S_{01} + S_{11} \cos [z \cos (ω_{c} t - θ_{1}) + φ_{1} (t)] \end{array}

\begin{array}{l} S_{2} (t) = S_{02} + S_{12} \cos {\frac{π β A}{V_{π}} \cos (ω_{c} t - θ_{2}) + \frac{2 π}{λ} [l_{2} + 2 l (t) + 2 d (t)]} \\ = S_{02} + S_{12} \cos [z \cos (ω_{c} t - θ_{2}) + φ_{2} (t)] \end{array}

where S₀_j and S₁_j (j = 1,2) are the amplitudes of the DC component and AC component, respectively, θ_j is the carrier phase delay, z = πβA/V_π is the sinusoidal phase modulation depth, l(t) is the displacement of the slowly moving stage in the common reference arm, and d(t) is the displacement to be measured. φ₁(t) = 2π[l₁ + 2l(t)]/λ and φ₂(t) = 2π[l₂ + 2l(t) + 2d(t)]/λ are phases to be demodulated corresponding to the monitor interferometer and probe interferometer, respectively. As the demodulated phase varies with the l(t), the quadrature components P₁(t) and P₂(t) in Eqs. (4) and (5) will vary periodically, with which the normalized parameters α₁ and α₂ can be always estimated by ellipse fitting of the Lissajous figure or calculating the amplitudes of the quadrature components. Then real-time normalization of the PGC demodulation is realized and the periodic nonlinearity can be reduced significantly. Subtracting the demodulated phases of the two interferometers, we can obtain

ϕ (t) = φ_{2} (t) - φ_{1} (t) = \frac{4 π}{λ} d (t) + \frac{2 π}{λ} (l_{2} - l_{1}) = \frac{4 π}{λ} d (t) + ϕ_{0} .

where ϕ₀ = 2π[l₂−l₁]/λ is the initial phase-difference between the monitor and probe interferometers. It can be seen that, the phases induced by l(t) are counteracted and the phase ϕ(t) is only related to the displacement d(t).

2.2 Real-time evaluation of the periodic nonlinearity using fixed-phase-difference detection method

According to Eqs. (4) and (5), without normalization of the quadrature components in PGC demodulation, the demodulated phase Φ(t) can be expressed as

Φ (t) = arc \tan (\frac{J_{1} (z) J_{2} (z^{'}) K_{1} \cos θ \sin φ (t)}{J_{1} (z^{'}) J_{2} (z) K_{2} \cos 2 θ \cos φ (t)}) = arc \tan (ν \tan φ (t))

where ν = [J₁(z) J₂(z′)K₁cosθ]/[J₁(z′) J₂(z)K₂cos2θ] is the proportional coefficient determined by the deviation of the theoretical phase modulation depth from the actual value, the performance of the multipliers and LPFs, and the carrier phase delay. Equation (14) can be expended and simplified into a linear component and a nonlinear component [23]:

Φ (t) = φ (t) + \frac{ν - 1}{2} \sin 2 φ (t)

It can be seen that, a nonlinear component with the period of π will be introduced if ν≠1, and the amplitude of the nonlinear component reaches its maximum of (v−1)/2 rad when φ(t) = kπ/2 + π/4, (k = 0,1,2…). Generally, the total harmonic distortion (THD) is used to evaluate the nonlinearity of PGC phase demodulation. With a single-frequency sinusoidal signal φ(t), the expansion of Eq. (15) is combined of the fundamental and harmonics of the phase signal, and the THD is defined as the ratio of equivalent root-mean-square (RMS) amplitude of all harmonics to the amplitude of the fundamental frequency. It is an effective method to evaluate the nonlinearity of phase demodulation. However, the phase signal is limited to single-frequency sinusoidal signal, which is inconvenient to implement in practical applications and is not appropriate to real-time evaluation. In addition, the value of THD is related to the initial phase and the dynamic range of the phase signal, and the error of phase demodulation cannot be reflected directly.

In order to realize real-time evaluation of the nonlinear phase error caused by demodulation, we use two photodetectors (PDs) placed at different probe points to detect the same interference signal. Supposing that the phase-difference induced by the interference fringe interval is δ, according to Eq. (15), the phase demodulated by another PD is

Φ^{'} (t) = φ (t) + δ + \frac{ν - 1}{2} \sin 2 [φ (t) + δ]

Then the phase-difference is

\begin{array}{l} Δ Φ (t) = Φ^{'} (t) - Φ (t) \\ = δ + \frac{ν - 1}{2} {\sin 2 [φ (t) + δ] - \sin 2 φ (t)} \\ = δ + (ν - 1) \sin δ \cdot \cos [2 φ (t) + δ] \end{array}

Equation (17) shows that the phase-difference ∆Φ comprises a fixed phase δ and a nonlinear component caused by v≠1, and the amplitude of the nonlinear component reaches its maximum of (v−1) rad when |sinδ| = 1 (δ = 90° or 270°). Therefore, for the modified EOM-based SPMI in Fig. 2, PD₂ and PD₃ are placed at an interval of quarter fringe to generate a fixed-phase-difference of 90° or 270° and a variation of the fixed-phase-difference can be detected if the nonlinear error is exist. As the common reference corner cube is mounted on a slowly moving stage, the phase-difference can be observed all the time during the measurement, thus the value of (v−1) can be obtained. The phase error caused by nonlinear demodulation can be deduced from the phase-difference ∆Φ no matter the target is in dynamic or static state.

3. Experiments and results

3.1 Experimental setup of the modified EOM-based SPMI

The experimental setup of the modified SPMI with an EOM was constructed as shown in Fig. 3. The laser source was a single-frequency He-Ne laser (XL80, Renishaw) with the wavelength of 632.990577 nm. The sinusoidal phase modulation was introduced by an EOM (EO-PM-NR-C1, Thorlabs) with lithium niobate crystal, and the displacement d(t) was provided by a nano-positioning stage (P-753.1CD, Physik Instrument) with a range of 15 μm and a resolution of 0.05 nm. The movement of the common reference corner cube l(t) is provided by another P-753.1CD stage, which moves back and forth with a speed of 100 nm/s in the range of 15 μm. The mechanical vibration induced by the moving stage with such a low speed can be neglected. The carrier signal and the detected interference signals were converted into digital signals by a 20 MHz simultaneous 4-chanel data acquisition card (PCI-9812, Adlink), and the real-time normalization and the periodic nonlinearity evaluation methods of PGC demodulation were processed in a computer based on Labview workbench.

Fig. 3 Experimental setup of the modified EOM-based SPMI.

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3.2 Experimental verification for real-time normalization in PGC demodulation

During the displacement measurement, the quadrature components of the two interferometers varied continuously with the moving of CC₁, and the normalization of the quadrature components is implemented automatically when either amplitude of the quadrature components has a deviation of 0.01 from 1 (i.e. the corresponding phase error is larger than 0.3°). An experiment was performed to verify the effectiveness of real-time normalization in PGC demodulation. In order to observe the effect of real-time normalization clearly, a discrepancy of 0.1 rad was set between the value of the phase demodulation depth used for signal processing and the actual value. CC₁ was moving with a speed of 100 nm/s, and CC₂ was in stationary. As shown in Fig. 4(a) and 4(c), φ₁(t) and φ₂(t) are demodulated phases of the monitor interferometer and the probe interferometer, respectively, and φ₂(t)-φ₁(t) is the phase-difference between them which is set as 200°. Before real-time normalization, the Lissajous figure of the quadrature components is an ellipse and the phase-difference varies periodically in 200 ± 10°. Figure 4(b) and 4(c) show that, after real-time normalization from the moment t₁, the Lissajous figure transforms into a unit circle and the phase-difference varies in 200 ± 1°. Figure 4(d) shows the corresponding demodulated displacements with CC₁ moving from 0 to 1.4 μm. d₁(t) and d₂(t) are demodulated displacements of the monitor and the probe interferometers, respectively, and d₂(t)−d₁(t) is the displacement difference between them. We can see that a periodic nonlinear error is added on the demodulated displacements of the two interferometers, and the displacement difference fluctuates in ± 10 nm before real-time normalization. After real-time normalization, the displacement difference is maintained to ± 1 nm.

Fig. 4 Results of real-time normalization verification. (a) Lissajous figure of P₁(t) and P₂(t). (b) Lissajous figure of Q₁(t) and Q₂(t). (c) Demodulated phases of the monitor and the probe interferometers. (d) Demodulated displacements of the monitor and the probe interferometers. The red dashed line is shifted by 50 nm to make the plots visible in (d).

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3.3 Experimental verification for real-time evaluation of the periodic nonlinearity

According to section 2.2, the periodic nonlinearity of PGC demodulation can be evaluated in real time by the fixed-phase-difference detection method and the optimal value of the phase-difference is 90° or 270°. In order to verify the effectiveness of the fixed-phase-difference detection method, the THD was also analyzed for the same demodulated displacement. Technically, the fixed-phase-difference detection method has no special requirement for the movement forms of the target, while in order to compare with the THD analysis, a sinusoidal signal is applied to the P-753.CD stage to make the probe corner cube CC2 has a sinusoidal movement with a frequency of 50 Hz and dynamic range of 10 rad. The two photodetectors of PD₂ and PD₃ in Fig. 3 were used to evaluate the periodic nonlinearity of PGC demodulation, and the phase-difference should be a fixed value around 90° in ideal demodulation. The phase-difference can be observed in real time. At the same time, the demodulated phase of PD₂ was analyzed to obtain the THD.

In the first experiment, so as to observe the phenomenon of nonlinear demodulation clearly, a discrepancy of 0.4 rad was set between the value of the phase demodulation depth used for signal processing and the actual value. The experimental results are shown in Fig. 5, in which φ₂(t) and φ₃(t) are demodulated phases of PD₂ and PD₃, respectively, and φ₂(t)−φ₃(t) is the phase-difference between them. Experimental results show that without real-time normalization, the phase-difference varies periodically in 90 ± 35° [Fig. 5(a)], and a periodic nonlinear error is added on the wrapped phase of φ₂(t) [Fig. 5(b)]. Figure 5(c) shows that the THD of the wrapped phase of φ₂(t) is 2.260%, and the signal-to-noise-and-distortion (SINAD) is 29.12 dB. In the second experiment, the quadrature components were normalized in real time and the experimental results are shown in Fig. 6. It can be seen that the phase-difference reduces to 90 ± 1° [Fig. 6(a)], and the wrapped phase of φ₂(t) becomes a relative smooth sinusoidal signal [Fig. 6(b)]. The corresponding THD is reduced to 0.933%, and the SINAD is improved to 40.40 dB as shown in Fig. 6(c).

Fig. 5 Results of PGC demodulation before real-time normalization. (a) Demodulated phases. (b) Wrapped phase of φ₂(t). (c) THD analysis of the wrapped phase of φ₂(t).

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Fig. 6 Results of PGC demodulation after real-time normalization. (a) Demodulated phases. (b) Wrapped phase of φ₂(t). (c) THD analysis of the wrapped phase of φ₂(t).

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3.4 Nanometer displacement measurement

Nanometer displacement measurement was performed to verify the method of real-time normalization in PGC demodulation. CC₁ was moving with a speed of 100 nm/s during the measurement and the quadrature components were normalized in real time. The nanometer displacement measurement was conducted by moving CC₂ with a step of 20 nm for 256 times (5.12 μm), and the measurement results are shown in Fig. 7. The displacement of the stage, the demodulated displacement and the deviation of the displacements between them are shown in Fig. 7(a). According to the measurement results, with real-time normalization, the average of the deviation is almost zero. The maximum error is 1.25 nm and the standard deviation is 0.56 nm. Figure 7(b) shows the FFT analysis of the displacement deviation shown in Fig. 7(a). According to Eq. (15), if a nonlinear error with the period of π occurs in the phase demodulation, there will be a peak at the second harmonic components of fringe. As shown in Fig. 7(b), no apparent peak emerges at the second harmonic components of fringe, and displacement measurement with nanometer accuracy can be realized without detectable nonlinearity beyond 0.1 nm in the modified EOM-based SPMI.

Fig. 7 Experimental results for nanometer displacement measurement. (a) Measurement results with the step of 20 nm in 5.12 μm. The red triangle line is shifted by 1 μm to make the plots visible. (b) FFT analysis of the displacement deviation.

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4. Conclusion

In this paper, real-time normalization in PGC demodulation has been realized in a modified EOM-based SPMI consisting of a monitor interferometer and a probe interferometer with a common slowly moving reference corner cube. In addition, a fixed-phase-difference detection method is proposed to evaluate periodic nonlinearity in PGC-Arctan demodulation. Several experiments were implemented for the feasibility verification of the proposed methods. The experimental results show that: (1) The phase error induced by the nonlinear demodulation can be significantly reduced by normalizing the quadrature components in real time. (2) With real-time normalization, the phase error caused by nonlinear demodulation is reduced to less than ± 1° and the corresponding THD is 0.933% with SINAD of 40.40 dB for the test sinusoidal movement with the frequency of 50 Hz and dynamical range of 10 rad. (3) Nanometer displacement measurement is realized with real-time normalization in PGC demodulation. Limited by the sampling rate of the data acquisition card and the data processing capability of the host computer, the measurement speed of the proposed SPMI in this paper is limited to several millimeters per second. If using high-speed data acquisition card with data processed by high performance FPGA, the proposed SPMI has a great potential in industrial applications such as machine-tool control and displacement sensor calibration.

Funding

National Natural Science Foundation of China (NSFC) (51527807, 51375461 and 51475435); Natural Science Foundation of Zhejiang Province (LZ18E050003); Program for Changjiang Scholars and Innovative Research Team in University (IRT_17R98).

Acknowledgments

Authors acknowledge the financial support from the 521 Talent Project and Science Foundation of Zhejiang Sci-Tech University.

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Real-time normalization and nonlinearity evaluation methods of the PGC-arctan demodulation in an EOM-based sinusoidal phase modulating interferometer

Abstract

1. Introduction

2. Theory

2.1 PGC-Arctan demodulation with real-time normalization

2.2 Real-time evaluation of the periodic nonlinearity using fixed-phase-difference detection method

3. Experiments and results

3.1 Experimental setup of the modified EOM-based SPMI

3.2 Experimental verification for real-time normalization in PGC demodulation

3.3 Experimental verification for real-time evaluation of the periodic nonlinearity

3.4 Nanometer displacement measurement

4. Conclusion

Funding

Acknowledgments

References and links

Cited By

Figures (7)

Equations (17)

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