Photonic generation of binary and quaternary phase-coded microwave signals by utilizing a dual-polarization dual-parallel Mach-Zehnder modulator

Peng Li; Lianshan Yan; Jia Ye; Yan Pan; Wei Pan; Bin Luo; Xihua Zou; Tao Zhou; Zhiyu Chen; Maowen Wang

doi:10.1364/OE.26.028013

1. Introduction

Phase-coded microwave signals have been widely used in modern radar and communication systems, due to their pulse compression capability. Conventionally, phase-coded microwave signals are generated in the electrical domain, which limits the operation frequency and time-bandwidth product of the system. Photonic methods to generate phase-coded microwave signals have attracted great attentions in the past few years, due to their advantages in term of small size, light weight, low insertion loss, immunity to electromagnetic interference and large time-bandwidth products [1–3].

So far, various photonic methods have been investigated to generate phase-coded microwave signals. In general, they can be divided into two categories: one is realized by optical spectral shaping followed by frequency-to-time mapping (FTTM) [4–7], the other is based on optical heterodyning [8–15]. In the first category, arbitrary waveform generation can be realized by utilizing an optical wave shaper together with a dispersive element. However, the system is bulk and not flexible for reconfiguration. In the second category, the basic idea is to heterodyne two phase-correlated optical wavelengths with different phase modulations. In order to separate the two phase-correlated optical wavelengths, an optical filter [8,9] or a length of polarization maintaining fiber (PMF) [10] is always used, which would reduce the flexibility of the system. To overcome this problem, phase-coded microwave signals can also be generated by utilizing different modulators, such as cascaded polarization modulators (PolMs), dual-polarization modulator, dual-polarization quadrature phase shift-keying (DP-QPSK) modulator, or dual-polarization dual-parallel Mach-Zehnder modulator (DP-DPMZM) [11–15].

The schemes discussed above are mostly used to generate binary phase-coded microwave signals. However, these methods are quite sensitive to the Doppler frequency shift. Moreover, compared with binary codes, polyphase codes exhibit better Doppler tolerance and sidelode characteristics [16,17]. In [18–20], the quaternary phase-coded microwave signals are generated by utilizing a phase modulator (PM) with 4-level drive signals. However, higher residual chirps may be introduced in the generated signals, which may reduce the tolerance against chromatic dispersion and nonlinear impairments [21].

In this paper, we propose and experimentally demonstrate a photonic method to generate binary and quaternary phase-coded microwave signals using a DP-DPMZM. The upper DPMZM driven by a radio frequency (RF) signal is used as an optical wavelength shifter, while the lower DPMZM driven by two independent electrical signals is used to generate a binary phase shift key (BPSK) or quadrature phase shift key (QPSK) signal. By combining the wavelength-shifted optical sideband and phase-modulated optical carrier, both binary and quaternary phase-coded microwave signals can be generated. In the proof-of-concept experiment, a 5-Gb/s phase-coded signal with the carrier frequency of 10 GHz and a 7.5-Gb/s phase-coded signal with the carrier frequency of 15 GHz are obtained.

2. Principle

Figure 1(a) shows the schematic diagram of the proposed binary and quaternary phase-coded microwave signals generation system, which consists of a LD, a DP-DPMZM, a polarizer, an EDFA and a PD. A linearly polarized optical wave from a LD is sent into a DP-DPMZM which is an integrated device consisting of a 90° polarization rotator (PR), two DPMZMs and a polarization beam combiner (PBC) [22]. The upper DPMZM consists of two push-pulled sub-MZMs and an optical phase shift as shown in Fig. 1(b), which can be used as an optical wavelength shifter. A RF signal from a microwave signal generator (MSG) is equally divided into two paths with 90° phase shift: one is applied to sub-MZM1, the other is applied to sub-MZM2. When the two sub-MZMs are biased at the minimum transmission point, the optical field at the output of the upper DPMZM can be expressed as

E_{up p e r} (t) = \frac{\sqrt{2}}{8} E_{0} \exp (j ω_{c} t) [\begin{array}{l} \exp (j m \cos (ω_{R} t) + j π) + \exp (- j m \cos (ω_{R} t)) \\ +exp (j φ) (\exp (j m \sin (ω_{R} t) + j π) + \exp (- j m \sin (ω_{R} t))) \end{array}]

where

E_{0}

and

ω_{c}

are the amplitude and angular frequency of the optical carrier,

m = π V_{R} / V_{π}

is the modulation index of the two sub-MZMs,

V_{R}

and

ω_{R}

are the amplitude and angular frequency of the RF signal,

V_{π}

is the half voltage of the sub-MZMs,

φ

is the phase difference between MZM1 and MZM2. Applying the Bessel function expansion to Eq. (1),

E_{up p e r} (t)

could be written as

E_{up p e r} (t) \approx \frac{\sqrt{2}}{8} E_{0} \exp (j ω_{c} t) [\begin{array}{l} - J_{0} (m) - j J_{1} (m) \exp (j ω_{R} t) - j J_{1} (m) \exp (- j ω_{R} t) \\ + J_{0} (m) - j J_{1} (m) \exp (j ω_{R} t) - j J_{1} (m) \exp (- j ω_{R} t) \\ - exp (j φ) (J_{0} (m) + J_{1} (m) \exp (j ω_{R} t) - J_{1} (m) \exp (- j ω_{R} t)) \\ +exp (j φ) (J_{0} (m) - J_{1} (m) \exp (j ω_{R} t) + J_{1} (m) \exp (- j ω_{R} t)) \end{array}]

where

J_{n}

is the nth-order Bessel function of the first kind. When

φ = π / 2

, Eq. (2) can be rewritten as

Fig. 1 (a) Schematic diagram of the proposed binary and quaternary phase-coded microwave signals generation scheme. The structure of the (b) U-DPMZM and (c) L-DPMZM. LD, laser diode; DP-DPMZM, dual-polarization dual-parallel Mach-Zehnder modulator; PC, polarization controller; Pol., polarizer; EDFA, erbium doped fiber amplifier; PD, photodetector; PR, polarization rotator; RF, radio frequency; PBC, polarization beam combiner.

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E_{up p e r} (t) \approx \frac{- \sqrt{2}}{2} j J_{1} (m) E_{0} \exp (j (ω_{c} + ω_{R}) t)

In the lower DPMZM, both the sub-MZMs are driven by the independent electrical coding signals generated from an arbitrary waveform generator (AWG) as shown in Fig. 1(c). When the two sub-MZMs are biased at the minimum transmission point, and the phase difference between them is set as $π / 2$ , the optical field at the output of the lower DPMZM can be written as

E_{l o w e r} (t) = \frac{\sqrt{2}}{8} E_{0} \exp (j ω_{c} t) [\begin{array}{l} \exp (j m_{1} s_{1} (t) + j π) + \exp (- j m_{1} s_{1} (t)) \\ +exp (j \frac{π}{2}) (\exp (j m_{2} s_{2} (t) + j π) + \exp (- j m_{2} s_{2} (t))) \end{array}]

where

s_{1} (t)

and

s_{2} (t)

are the polar binary coded data (i.e. + 1, −1),

m_{1} = π V_{1} / V_{π}

and

m_{2} = π V_{2} / V_{π}

are the modulation index of MZM3 and MZM4 respectively,

V_{1}

and

V_{2}

are the amplitudes of the electrical coding signals. When

V_{1} = V_{2}

, Eq. (4) can be rewritten as

E_{l o w e r} (t) = {\begin{cases} \frac{1}{2} E_{0} | \sin (m_{1}) | \exp j (ω_{c} t - \frac{π}{4}) (s_{1} (t) = 1, s_{2} (t) = 1) \\ \frac{1}{2} E_{0} | \sin (m_{1}) | \exp j (ω_{c} t + \frac{π}{4}) (s_{1} (t) = 1, s_{2} (t) = - 1) \\ \frac{1}{2} E_{0} | \sin (m_{1}) | \exp j (ω_{c} t + \frac{3 π}{4}) (s_{1} (t) = - 1, s_{2} (t) = 1) \\ \frac{1}{2} E_{0} | \sin (m_{1}) | \exp j (ω_{c} t - \frac{3 π}{4}) (s_{1} (t) = - 1, s_{2} (t) = - 1) \end{cases}

As can be seen from Eq. (5), quaternary phases (i.e. −45°, 45°, −135°, 135°) have been generated by properly setting the two polar binary coded data. Then the modulated signals from the two DPMZMs are combined by a PBC with a 90° PR employed at one input port. The output signals are sent to a polarizer with its principal axis oriented an angle of 45° to one principal axis of DP-DPMZM. The optical field at the output of the polarizer can be expressed as

E_{P o l .} = \frac{\sqrt{2}}{2} E_{up p e r} (t) + \frac{\sqrt{2}}{2} E_{l o w e r} (t)

When the output optical signal is sent to a square-law PD, we can obtain

\begin{matrix} i (t) = R E_{P o l .} (t) * E_{P o l .} (^{t) *} \\ = {\begin{cases} \frac{\sqrt{2}}{4} R J_{1} (m) E_{0}^{2} | \sin (m_{1}) | \cos (ω_{R} t - \frac{π}{4}) (s_{1} (t) = 1, s_{2} (t) = 1) \\ \frac{\sqrt{2}}{4} R J_{1} (m) E_{0}^{2} | \sin (m_{1}) | \cos (ω_{R} t + \frac{π}{4}) (s_{1} (t) = 1, s_{2} (t) = - 1) \\ \frac{\sqrt{2}}{4} R J_{1} (m) E_{0}^{2} | \sin (m_{1}) | \cos (ω_{R} t + \frac{3 π}{4}) (s_{1} (t) = - 1, s_{2} (t) = 1) \\ \frac{\sqrt{2}}{4} R J_{1} (m) E_{0}^{2} | \sin (m_{1}) | \cos (ω_{R} t - \frac{3 π}{4}) (s_{1} (t) = - 1, s_{2} (t) = - 1) \end{cases} \end{matrix}

where R is the responsivity of the PD. As can be seen from Eq. (7), a quaternary phase-coded microwave signal with the carrier frequency of

ω_{R}

is generated.

3. Experimental setup and results

To verify the proposed scheme, we build an experimental setup as shown in Fig. 1. A continuous optical wave generated from a tunable laser source (TLS, Yenista optics) with the center wavelength of 1549.3nm and power of 11 dBm is sent to a DP-DPMZM (FUJITSU, FTM7977). A RF signal with the frequency of 10 GHz and power of 15 dBm generated from a MSG (Anritsu, MS2840A) is applied to the two branches of the upper DPMZMM with a 90° electrical hybrid between them. The two electrical coding signals generated from an AWG (M8195A) are amplified by an electrical amplifier (EA, SHF 100 AP) and then sent to the lower DPMZM. The output of the DP-DPMZM is sent to a polarizer via a polarization controller (PC). The optical signal at the output of the polarizer is amplified by an EDFA (Amonics) and then detected by a PD (Agilent 11982A) with the 3-dB bandwidth of 15 GHz. The current at the output of PD is monitored by a digital storage oscilloscope (Keysight, DSOZ634A) with the bandwidth of 65 GHz and the sampling rate of 160 GSa/s. The optical spectra are measured by an optical spectrum analyzer (OSA, Yokogawa, AQ6370D) with the resolution of 0.02 nm.

Firstly, the frequency of the upper DPMZM-driving RF signal is set to be 10 GHz. Two independent electrical encoding signals with the rate of 5 Gbit/s are fed to the lower DPMZM. The measured optical spectrum at the output of the upper DPMZM is depicted in Fig. 2(a). As can be seen that the + 1-order sideband is dominant, while the optical carrier and −1-order sideband are suppressed 10.6 dB and 33.2 dB respectively. Figure 2(b) shows the eye diagram at the output of the lower DPMZM measured by an optical sampling oscilloscope (Agilent, Infiniium, 86100C). A double intensity dip indicates the characteristic for QPSK modulation, and the period of the generated QPSK signal is measured to be 200 ps.

Fig. 2 (a) The optical spectrum at the output of the upper DPMZM and (b) The eye diagram at the output of the lower DPMZM.

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At the output of the PD, the binary and quaternary phase-coded microwave signals are generated. Figures 3(a) and 3(b) show the electrical spectra of the generated binary and quaternary phase-coded microwave signal, respectively. Figures 3(c) and 3(d) show the generated binary phase-coded microwave signal and extracted phase shift information corresponding to the encoding signal based on coherent demodulation in the duration of 6.4 ns. Two 32-bit binary signals are generated by the AWG with the same pattern of “1 1 1 0 0 0 1 0 1 0 1 1 0 0 1 0 1 1 1 0 1 0 0 0 1 1 1 1 0 1 0 1” (bit ‘1’ stands for polar binary ‘ + 1’, bit ‘0’ stands for ‘-1’). As can be seen that a phase shift of ~180° is observed between the symbols ‘00’ and ‘11’, which agrees well with the theoretical analysis. When we adjust one of the 32-bit binary signals to the pattern of “0 1 1 0 0 1 0 1 1 1 0 0 1 1 0 0 1 1 0 1 0 0 1 1 1 0 0 0 0 1 0 1”, the generated quaternary phase-coded microwave signal and extracted phase shift information corresponding to the encoding signal are shown in Figs. 3(e) and 3(f). As can be seen that the phase shift have four levels. Taking, for example, ‘10’ as a reference, then the symbols ‘11’, ‘01’ and ‘00’ are corresponding to a phase difference ~90°, ~180°, and ~270° respectively.

Fig. 3 Electrical spectral of the generated 10 GHz (a) binary and (b) quaternary phase-coded signal. (c) Waveform of the generated 10 GHz binary phase-coded microwave signal, and (d) Extracted phase shift information from (c). (e) Waveform of generated 10 GHz quaternary phase-coded microwave signal, and (f) Extracted phase shift information from (e).

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In order to implement the pulse compression of the generated binary and quaternary phase-coded microwave signals, a matched filter with the frequency response equals the complex conjugate of the signal is designed. The output of the filter is the inverse Fourier transform of the product of the generated signal spectrum and the matched filter response. The autocorrelation curves of the simulated binary and quaternary phase-coded microwave signal are shown in Figs. 4(a) and 4(c), respectively. The peak-to-sidelobe ratios (PSRs) are 6.4 dB and 5.8 dB and the corresponding pulse compression ratios (PCRs) are 64 and 62 respectively. Figures 4(b) and 4(d) show the autocorrelation curves of the measured binary and quaternary phase-coded microwave signal. The PSRs are 5.31 dB and 5.14 dB, and the PCRs are 60 and 57. The measured results agree well with the simulated ones.

Fig. 4 The autocorrelation of the binary phase-coded microwave signal: (a) simulated results and (b) experimental results, and the quaternary phase-coded microwave signal (c) simulated results and (d) experimental results.

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To verify the frequency tunability of the system, we adjust the frequency of the upper DPMZM-driving RF signal to be 15 GHz and set the rate of the two electrical encoding signals to be 7.5 Gbit/s. Figures 5(a) and 5(b) show the generated binary phase-coded microwave signal and corresponding phase shift information. As can be seen that the phase shift of the ~180° between the symbol ‘00’ and ‘11’. Figures 5(c) and 5(d) show the generated quaternary phase-coded microwave signal and corresponding phase shift generated information. It can be seen that the phase shift have four levels (i.e. −45°, 45°, −135°, 135°). The pulse compression capability is also studied as shown in Fig. 6. The PSRs are 4.94 dB and 4.73 dB and PCRs are 70 and 65.

Fig. 5 (a) Waveform of generated 15 GHz binary phase-coded microwave signal, and (b) Extracted phase shift information from (a); (c) Waveform of generated 15 GHz quaternary phase-coded microwave signal, and (d) Extracted phase shift information form (b).

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Fig. 6 The autocorrelation of the generated (a) binary phase-coded microwave signal and (b) quaternary phase-coded microwave signal.

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4. Conclusion

In conclusion, a method to generate binary and quaternary phase-coded microwave signals has been proposed and demonstrated. Theoretical analysis shows that the proposed scheme has a wide operation frequency range since no optical or electrical filters are used. The only limitation is the bandwidth of the DP-DPMZM and PD. Experimental results show that a10-GHz RF signal with the coding rate of 5 Gb/s and a 15-GHz RF signal with the coding rate of 7.5 Gb/s are obtained. Their autocorrelation results show a good pulse compression capability. The proposed scheme has a compact structure, wide operation bandwidth and great tolerance to the chromatic dispersion and nonlinearities, which can be used in modern radar systems.

Funding

National Basic Research Program of China (2013CBA01704); National Natural Science Foundation of China (NSFC) (61335005, 61771438, 61860206006); Ministry of Education United Foundation of Equipment Pre-Research (6141A020334).

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Photonic generation of binary and quaternary phase-coded microwave signals by utilizing a dual-polarization dual-parallel Mach-Zehnder modulator

Abstract

1. Introduction

2. Principle

3. Experimental setup and results

4. Conclusion

Funding

References and links

Cited By

Figures (6)

Equations (7)

Optics Express