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The generation of stable and high-quality single-sideband (SSB) multi-carrier source based on recirculating frequency shifter (RFS) is analyzed theoretically and realized experimentally. The impact factors originated from the modulator intrinsic imperfections, deviation from the right operation bias voltage, as well as the unbalanced amplitude and phase of the radio frequency (RF) drive signals, have different influences on the output spectrum of the transfer function, which is the decisive factor in generating the high-quality multi-carrier output. Based on the theoretical analysis, the stable and high-quality 50-tone output was successfully realized. The experiments under some implementation imperfections have also been carried out. The imperfect and low-quality output results are in good agreement with the theoretical analysis.
©2011 Optical Society of America
1. Introduction
In order to achieve the T-bit/s high-speed optical transmission for the increasing demand of data traffic, there are some technologies such as coherent wavelength-division-multiplexing (CoWDM) [1–3], dense-wavelength-multiplexing (DWDM) with polarization-division-multiplexing (PDM) [4–6], and especially coherent optical orthogonal-frequency-division-multiplexing (CO-OFDM) with different modulation formats [7–17] have been investigated widely in recent years. To further increase the spectral efficiency of transmission system, the frequency-locked multi-carrier source is needed. There are some methods to generate these carriers, including the typical cascaded Mach-Zehnder modulators (MZMs) [18], optical frequency comb source based on a sine wave driven discrete mode (DM) laser [19], and more recently, the single-side-band (SSB) modulator based on recirculating frequency shifter (RFS) [2,3,12–17], which has been investigated theoretically and experimentally. Compared with the three technologies mentioned above, the last one has the advantages of lower drive voltage and the ability to generate larger number of carriers. In addition, these carriers are exactly frequency-locked and far less sensitive to phase noise and non-linear propagation effects. By applying this method, the 24-tone [14–16] and 36-tone [12,13] multi-carrier sources have been generated successfully. Moreover, the crosstalk and stability of the multi-carrier generator under the perfect operational condition have also been investigated in [20] and [21]. Actually, there are many impact factors to affect the quality of the multi-carrier output, such as the imbalance of I/Q modulator (IQM) and radio-frequency (RF) drive signals in practice. Although the imbalance factors of IQM used as a modulator in the OFDM transmitter has been analyzed in [22], the impact on the multi-carrier source which has a significant difference has not been studied in the previous literature. In order to achieve a relative high-quality multi-carrier output under the actual imperfect implements, the analysis of the impact factors is necessary and would give some useful results to carry out the next Tb/s optical transmission.
In this paper, we theoretically and experimentally analyze the influences due to the impact factors, which are originated from I/Q modulator and RF drive signals on the output quality of the SSB modulator based RFS. These impact factors include the intrinsic imbalance of modulator, the imbalance of operation bias voltage, and the amplitude and phase imbalance of RF drive signals. A crosstalk coefficient used to quantify the impact of the all impact factors on the output spectrum has been defined. The simulation results have been obtained with a set of given parameters. According to the theoretical analysis, a stable and flat 50-tone high-quality SSB multi-carrier experiment has been demonstrated. Moreover, to verify the theoretical analysis, the experiments of the impact of these impact factors on the output of multi-carrier generator have also been carried out. The experimental results are in good agreement with the theoretical analysis results.
2. Theoretical analysis of the impact factors
The schematic of SSB modulator based RFS used as a multi-carrier source is shown in Fig. 1(a) . The configuration is composed of a closed fiber loop, which consists of a 50:50 coupler, an IQM, a tunable optical band-pass filter (BF) used to control the number of desired carriers, a polarization controller (PC) and an optical amplifier (OA) which is used to compensate the modulation losses in the loop. The IQM is driven with two equal-amplitude but π/2 phase shifted RF clock signals through I and Q ports, to induce a positive (or negative) frequency shifting to the input signal from the tunable laser source (TLS) which acts as a seed signal. The IQM which is commercially available in an integrated form is illustrated in Fig. 1(b). Both the sub-Mach-Zehnder Modulators (sub-MZMs) are operated in the push-pull mode. The operating signals of sub-MZM1, sub-MZM2 and phase modulator (PM) are denoted by S I(t), S Q(t) and S PM(t) respectively. The phase shifts ϕ I1, ϕ I2, ϕ Q1, ϕ Q2 and ϕ PM correspond to the upper and lower arms of sub-MZMs and PM, which are induced by corresponding voltages respectively.
With the consideration of the actual operational conditions, there are another three kinds of impact factors to affect the quality of the multi-carrier generator except the nonlinearity of IQM which has been analyzed in [21]. These factors include (i) intrinsic imperfections due to manufacture (include the PM insertion loss δ and the voltage amplitude deviations ΔV I and ΔV Q which come from the upper and lower arms of each sub-MZM), (ii) the deviation of bias operation voltage, and (iii) the imbalance of RF clock drive signal. We represent the input signal as E in(t) = Aexp(j2πf 0t), operation drive signals as S I(t) = V bI + V ppIcos(2πf mt), S Q(t) = V bQ + V ppQsin(2πf mt + Δθ), and S PM(t) = V bPM. Assuming a positive frequency shifting process with N + 1 desired carriers, the generated carriers of the multi-carrier generator can be denoted as f 0, f 1, …, f N. Namely, the desired signal after the first round trip (RT) is f 1 whose frequency equals to f 0 + f m, and f N with its frequency equals to f 0 + Nf m. By considering these impact factors, the normalized transfer function which is defined as the output of multi-carrier source after the first RT can be given by
where ϕ RT is the phase delay per RT. The phase differences induced by the phase modulators (PMs) for different arms are shown as followswhere V πI, V πQ, V πPM represent the half-wave voltage of Sub-MZM1, Sub-MZM2 and PM respectively. ΔV bI, ΔV bQ, ΔV bPM represent the deviation of DC bias voltage from the right bias operation point, and ΔV m is the amplitude imbalance between I and Q ports RF drive signals (so V ppQ = V ppI + ΔV m). In Eq. (2), we can see that there are so many impact factors to affect the output of IQM. Therefore, to achieve a good quality of multi-carrier output, we should apply the parameters properly in practice.2.1 The analysis of the impact of the intrinsic imperfections due to manufacture
At first, we only analyze the impact of the intrinsic imperfections due to manufacture of IQM on the transfer function, while the other factors are assumed to operate at the perfect conditions. Namely, the main parameters δ, ΔV I, ΔV Q are analyzed and the other parameters such as ΔV bI, ΔV bQ, ΔV bPM, ΔV m and Δθ are all equal to zero. Neglecting all the harmonics beyond 3rd-order, then Eq. (1) can be expressed by the Jacobi-Anger expansion with substituting the phase shifts in Eq. (2)
where the coefficients of the 0th - to 3rd - order harmonics are as followswhere δ mI = (πV ppI)/(2V πI) = δ mQ = (πV ppQ)/(2V πQ) denote the phase modulation depth; Δδ mI = (πΔV I)/(4VπI), Δδ mQ = (πΔV Q)/(4V πQ) denote the deviation of phase modulation due to intrinsic imperfections of IQM, and Jk(.) (k = 0, 1, 2, 3) are the first kind Bessel functions respectively. From Eq. (3) and (4), we can see that the output spectrum after the 1st RT will generate the 0th to 3rd order harmonics which contain all the unwanted crosstalk components f 0, f -1, f ± 2, and f ± 3. These crosstalk components will affect the final output of SSB modulator. Therefore, we must lower the power of all the undesired tones to make the power difference between the desired signal and undesired tones as lager as possible. However, there will be still a dominant crosstalk component whose frequency (denoted by f c briefly, and will be used in the following sections) will be higher or lower than the desired signal f 1, and then affect the final output quality. We can analyze the property of these harmonics in frequency- domain by using Eq. (4). Taking into account the symmetric property of the harmonics in frequency-domain, we can just need to analyze the property of crosstalk component with its frequency higher or lower than f 1 after the first RT. Here, we choose the crosstalk components with their frequencies lower than f 1, namely, the crosstalk components will contain fc = f 0, f -1, f -2 and f -3. This choice will also be used and convinced to be appropriate in following sections.Figure 2 shows the impact of the intrinsic imperfections on the output transfer function theoretically. In the simulation, we assume that V πI = V πQ, and the amplitude of RF drive signal V m is 0.36V πI, which is equal to the optimum value obtained in [21]. ΔP i (i = 0, 1, 2, 3) = Pf c – Pf 1 represents the normalized power difference between the crosstalk component and desired signal. For instance, ΔP 0 = Pf 0 – Pf 1. In Fig. 2(a), we find that only the odd-order harmonics crosstalk appears when ΔV I = ΔV Q = 0 which corresponds to the situation of the two arms of each sub-MZM which are balanced. In addition, the symmetric increasing trend as the function of intrinsic loss δ is shown for the crosstalk component f -1. Compared with Fig. 2(b) and (c), this symmetric increasing trend vs. absolute of δ is maintained while the output spectrum will not contain f -1 as long as (ΔV I /V πI) = (ΔV Q /V πQ). However, the even-order crosstalk components, especially for f 0, begin to be the dominant factor to affect the output spectrum under the case of less insertion loss, which is the actual operational condition. When (ΔV I /V πI) ≠ (ΔV Q /V πQ), the component of f -1 will appear (Fig. 2(c)). With the variation of δ from −1 to 1, the crosstalk components f -2 and f -3 keep increasing trends but can be neglected within a little range of δ. To validate the theoretical analysis, the simulation results are shown in the bottom line of Fig. 2 corresponding to the cases which are shown in the top line of Fig. 2. In these figures, the horizontal axis is labeled with λ n = λ 0 + n ∆λ, corresponding to f -n = f 0 – n ∆f, where λ 0 = c / f 0, ∆λ = c ∆f /(f 0)2 (c is the speed of light). This label will also be used in following sections. The conditions corresponding to the case 1 ~case 3 are ΔV I /V πI = ΔV Q /V πQ = 0, ΔV I /V πI = ΔV Q /V πQ = −0.1, and ΔV I /V πI = −0.1, ΔV Q /V πQ = −0.3 under the same δ = 0 respectively. According to these figures, we can see that the trends of the crosstalk components change are in good agreement with the above theoretical analysis.
Fig. 2 The impact of intrinsic imperfection on the output of transfer function with (a) ΔV I /V πI = ΔV Q /V πQ = 0; (b) ΔV I /V πI = ΔV Q /V πQ = −0.1; (c) ΔV I /V πI = −0.1, ΔV Q /V πQ = −0.3.
2.2 The analysis of the impact of bias operation voltage
Here, we assume that the IQM is perfectly balanced and RF drive signals are balanced. Namely, the parameters that we discussed in this section are ΔV bI, ΔV bQ, ΔV bPM, with all the other parameters being equal to zero. Under this condition, we obtain the transfer function followed the same way in section 2.1 as follows
where the coefficients of the 0th - to 3rd - order harmonics are as followswhere Δδ bI = (πΔV bI)/(2V πI), Δδ bQ = (πΔV bQ)/(2V πQ), and Δδ PM = (πΔV bPM)/(2V πPM) represent the bias voltage deviations from the right operation voltage respectively. The parameters δ mI and δ mQ are the same as defined in section 2.1. From Eq. (5) and (6), we can see clearly that the even-order harmonics appear as long as the bias voltage deviation is not equal to zero. Obviously, to reduce the harmonics crosstalk, we should apply the bias voltage at the right operation voltage as accurate as possible in practice.The impact of the deviation ΔV bPM from right bias operation voltage of PM on the output transfer function is shown in the top and bottom lines in Fig. 3 . With the increase of ΔV bPM from −V πPM to V πPM, we can see that, (i) the crosstalk component f 0 keeps in decreasing trend and is the dominant component if ΔV bI and ΔV bQ are not equal to zero; (ii) f -1 firstly decreases to the minimum value at ΔV bPM = 0 and then increases with a symmetric property; (iii) f -2 keeps increasing, but f -3 keeps a relative fix value in all cases. In addition, there will be the 1st-order harmonics crosstalk when (ΔV bI/ V πI) ≠ (ΔV bQ/ V πQ) even if ΔV bPM = 0. The simulation results are shown in the bottom line of Fig. 3 corresponding to the cases which are shown in the top line of Fig. 3. The parameters in these figures are: (i) case 1, ΔV bI /V πI = ΔV bQ /V πQ = −0.1, ΔV bPM/ V πPM = 0; (ii) case 2, ΔV bI /V πI = −0.1; ΔV bQ /V πQ = −0.3, ΔV bPM/ V πPM = 0; (iii) case 3, ΔV bI /V πI = ΔV bQ /V πQ = −0.1, ΔV bPM/ V πPM = −0.2; and (iv) case 4, ΔV bI /V πI = −0.1; ΔV bQ /V πQ = −0.3, ΔV bPM/ V πPM = −0.2 respectively. Obviously, the trends of the crosstalk components are also in good agreement the above theoretical analysis.
When ΔV bPM is at a certain value, the impact curves are shown in Fig. 4(a)–(d) . From (a) and (b), we can see that all crosstalk components give the symmetric properties with the variation of ΔV bI. Further, to obtain the acceptable deviation value from Fig. 4(a) and (b), the theoretical curves within a small range of [−0.1 0.1]V πI are shown in Fig. 4(e) and (f) corresponding to these detailed range denoted by the case A and B respectively. In case A, f 0 will be the dominant crosstalk component as long as |ΔV bI| > 0.005V πI. When |ΔV bI| < 0.005V πI, f -3 will be the dominant crosstalk component. In addition, to keep the power difference of the dominant crosstalk lower than desired signal around 30dB, the acceptable range of ΔV bI is only from −0.012V πI to 0.012V πI. With the consideration of the symmetric property, ΔV bQ will be also within in the range of [−0.012 0.012]V πQ when ΔV bI = 0. However, when ΔV bQ is not equal to zero at arbitrary value of ΔV bPM, the f 0 will be the dominant crosstalk component as shown in (c) and (d). In the view of actual implementation, this case will be unacceptable.
2.3 Analyzing the impact of the imbalance of RF drive signals
In this case, we focus on the impact of the voltage amplitude and phase differences between I-port and Q-port RF drive signals on the output. Assuming the IQM is perfectly balanced and all modulators are operating at their right points, namely, only the parameters of ∆θ and ∆V m are considered. Then the transfer function is expressed as follows
where the coefficients of the 1st – order and 3rd - order harmonics are as followswhere δ m = δ mI = δ mQ, Δδ m = (π∆V m)/(2Vπ), C 1 = (δ m + Δδ m) cos(Δθ), C 2 = (δ m + Δδ m) sin(Δθ).Evidently, we can see easily from Eq. (7) and (8) that there are no even-order harmonics even though the RF drive signals are not perfectly balanced. This result is different from the above two situations discussed in Section 2.1 and 2.2.
The impact of ∆θ and ∆V m of RF drive signals on the output transfer function is shown in Fig. 5 . In (a), we can see that there are only the odd-order harmonics in the output spectrum and the power of f -1 and f -3 shows the symmetric variation property vs. the variation of phase deviation ∆θ when ∆V m = 0. This symmetric property can be deduced from Eq. (8) easily. However, the output property of the crosstalk component with the variation of amplitude deviation ∆V m is asymmetric as shown in Fig. 5(b) when ∆θ = 0. We can see that the power of f -1 changes much faster when ∆V m<0 than that when ∆V m>0. Compared to the above two imbalance situations in Section 2.1 and 2.2, the significant difference here is that the even-order harmonics crosstalk will not appear under the imbalance of two RF signals which can be convinced by Eq. (7). As shown in Fig. 5(c) and (d), to keep the power differences of dominant crosstalk components lower than desired signal f 1 round 30dB, ∆θ and ∆V m should be within the range of [−0.02 0.02]π and [−0.024 0.026]V ppI respectively.
Fig. 5 The impact of amplitude and phase imbalance of RF drive signals on the output of transfer function.
Figure 6 shows the simulation results that are corresponding to the different cases which are shown in Fig. 5. The parameters in these figures are: (i) case 1, ΔV m = 0, Δθ = −0.1π; (ii) case 2, ΔV m = 0, Δθ = −0.33π; (iii) case 3, ΔV m = 0, Δθ = −0.5π; and (iv) case 4, Δθ = 0, ΔV m = −0.2V PPI; (v) case 5, Δθ = 0, ΔV m = −0.72V PPI; and (vi) case 6, Δθ = 0, ΔV m = −0.8V PPI, respectively. Take case 2 for example, in which ∆θ = −0.33π, the component of f -1 is just a slight lower than of f 1, while f -3 has the lowest value. This simulation result is in good agreement with the above theoretical analysis (Fig. 5(a)). Concurrently, the other simulation results are also coincident with the theoretical analysis under the other different cases.
Fig. 6 The impact of amplitude and phase imbalance of RF drive signals on the output of transfer function with case 1 ~6 in Fig. 5(a) and (b) respectively.
2.4 Analysis of the output stability due to the impact factors
Based on the above analysis, the frequency of the dominant crosstalk component will be lower than f 1 in most cases (see the bottom line of Fig. 2 and 3, Fig. 6). Therefore we will only consider the crosstalk components of f c < f 1. To analyze the impact of the crosstalk components on the final output spectrum, the normalized transfer function shown as in Eq. (1) can be rewritten simply as follows
where b and n 0 ( = −3, −2, −1, 0) are the normalized coefficient and the order of crosstalk component respectively.Under the basic assumption of the first order approximation, and the optical filter inside the RFS can block any frequency less than or equal to the seed frequency f 0 and higher than f N, so the final output of SSB multi-carrier is obtained as follows when f c < f 1
where the normalized crosstalk coefficient C n can be expressed as followsObviously, we can see from Eq. (11) that the crosstalk components can reach the stable state only after the (N-n 0 + 1)-th RT when f c < f 1. This conclusion has a good agreement with [21] when the crosstalk components originated from the 3rd-order harmonics without considering the imbalance impact factors. In addition, the amplitude of crosstalk components is different for the different dominant harmonics labeled by n. The worst-case crosstalk values C max are |Nb|, |(N−1)b| and |(N−2)b| respectively corresponding to the dominant crosstalk component of the f 0, f- 1 and f -2 harmonics.
The effective OSNR of the multi-carrier source, under the three imperfection situations discussed in Section 2.1 ~2.3, can be obtained with the similar analysis in [21]. The effective OSNR from OA and the dominant crosstalk components can be expressed as follows respectively
where P out is the saturation power of the OA, NF is the noise figure, and L total is the total loss containing the losses of coupler, filter, IQM insertion and others. So the effective OSNR of the multi-carrier source can be obtained by combining the above two equations as followsAssuming all the parameters P out, NF, L total, N and |b| are the same, we can see from Eq. (12)–(14) that the effective OSNR for the different cases will be little different even if the dominant crosstalk component originated from different harmonics (N is large and fixed, n 0 is small and different).
According to the above analysis, the f 0 and f -1 will be the dominant crosstalk components to affect the entire multi-carrier output quality under the implementation imperfections. To study their influence on the entire output, the simulation results are shown in Fig. 7 and Fig. 8 respectively. In Fig. 7(a), f 0 (λ 0 in wavelength) is the dominant crosstalk component whose power is about 25dB lower than f 1. With the assumption of the total desired carriers number is 50, we can see from Fig. 7(b) that the last tone will undergo the maximum crosstalk in the final output spectrum of multi-carrier source. However, when the f -1 (λ 1) turns out to be the dominant harmonic as shown in Fig. 8(a), the last two tones will undergo the maximum crosstalk from Fig. 8(b). Again, these simulation results are in good agreement with the Eq. (10) and (11).
Fig. 7 The simulation output of the 50-tone multicarrier output when f 0 is the dominant harmonic. (a) the transfer function output; (b) the 50-tone output.
Fig. 8 The simulation output of the 50-tone multicarrier output when f -1 is the dominant harmonic. (a) the transfer function output; (b) the 50-tone output.
3. Experimental setup and results
To further study the impact of the impact factors on the multi-carrier output, the experimental setup is implemented as shown in Fig. 1(a). The TLS wavelength runs at λ 0 = 1549.9nm (f 0 = 193.425THz), the bandwidth of optical band-pass filter is set to 5nm, and the frequency of clock drive signal is 12.5GHz. In addition, the saturation output power of Raman amplifier is 27dBm, and the insertion loss of our IQM, filter and coupler are about 13dB, 3dB, and 4dB respectively. In order to achieve the good performance, we use two microwave power amplifiers to amplify the amplitude of I- and Q- ports RF drive signals independently, and use two power attenuators to make them be balanced.
At first, we obtain the best output of transfer function by applying the proper operation condition as shown in Fig. 9(a) . Due to the intrinsic imbalance of our IQM (un-adjustable factor), the maximum difference between f 1 (λ -1) and f 0 (λ 0) is around 25dB. Under the appropriate adjusting, the flat and stable 50-tone multi-carrier output with the spacing of 12.5GHz, which the total bandwidth is 5nm, is successfully achieved as shown in Fig. 9(b). We can see that the last tone (as marked with f 49 in the red circle) indeed undergoes the maximum crosstalk as analyzed in Section 2.4 and shown in Fig. 7(b), and the effective OSNR of the last tone is around 20dB.
Fig. 9 The experiment results under the balanced operation condition for (a) the transfer function output and (b) the 50-tone output.
Then the impact of the deviation of DC bias from the right operation point on the multi-carrier source output is shown in Fig. 10 . The parameters in the figure are as follows, (a) ∆V bI = 0.1V; (b) ∆V bI = 0.3V; (c) ∆V bI = 0.5V; (d) ∆V bI = −0.2V; and (e) ∆V bI = −0.4V; (f) ∆V bI = −0.6V. Insert (g) shows the experimental data of the maximum output power difference among the carriers with the variation of ∆V bI. Obviously, this experimental result shows the approximately symmetric property which is again in a good agreement with the above analysis in Fig. 4(a).
Fig. 10 The output spectrum with ∆V bI = :(a) 0.1V; (b) 0.3V; (c) 0.7V; (d) −0.1V; (e) −0.2V; (f) −0.4V.
Subsequently, the different output spectrums are obtained by changing the amplitude difference of RF drive signals of I and Q ports as shown in Fig. 11 . The parameters in these figures are as follows, (a) ∆P m = 0.5dB; (b) ∆P m = 1dB; (c) ∆P m = 1.5dB; and (d) ∆P m = −0.5dB; (e) ∆P m = −1dB; (f) ∆P m = −1.5dB respectively. With the comparison of the figures in the top and bottom lines in Fig. 12 , we can see that the entire output deteriorates much faster with the increasing of ∆P m in the range of negative (∆V m < 0) than that in positive values (∆V m > 0). This experimental result is still in good agreement with the theoretical analysis in Fig. 5(b).
Fig. 11 The output spectrum with ∆P m = (a) 0.5dB; (b) 1dB; (c) 1.5dB; (d) −0.5dB; (e) −1dB; (f) −1.5dB.
Figure 12 shows the impact of the phase deviation ∆θ on the entire output of the multi-carrier source. The parameters used in the figure are as follows, (a) ∆θ = −π/4; (b) ∆θ = −π/2; (c) ∆θ = π/4; (d) ∆θ = π/2. We can see that the output performance goes worse with the increasing of ∆θ. However, the variation of ∆θ will be within the period of π with considering the periodicity of trigonometric function. Furthermore, the output of multi-carrier generated indeed shows the symmetric property just as shown in Fig. 5(a).
Based on the above theoretical and experimental analysis, to mitigate these imperfection influences, we can apply the tunable power attenuator to balance the amplitude of the two RF drive signals, adjust time delay on the signal pattern generator to make their time synchronously, and use more accurate DC voltages to improve the bias voltage of the modulators. In addition, to achieve a high-quality multi-carrier source based on RFS, there are some other factors to be taken into account such as the external cavity laser (ECL) source with more stable frequency and phase, optical amplifier with lower noise figure (NF) and IQM with lower insertion loss, besides the above implementation imperfections.
5. Conclusions
Guided by the theoretical analysis, the stable and high-quality 50-tone multi-carrier was successfully achieved experimentally. The impacts of the crosstalk component induced by the impact factors originated from the intrinsic imperfections of the IQM, the deviation of DC bias voltage from the right operation point, and the imbalance of the amplitude and phase of the RF drive signals on the output of SSB-based RFS multi-carrier generator have been analyzed both theoretically and experimentally. In the positive frequency shifting process, with the effort of adjusting the appropriate parameters as possible as we can, the f 0 will be the dominant crosstalk component under the intrinsic imperfections of IQM (which we cannot control) and DC bias imbalances. On the other hand, f -1 will be just the dominant crosstalk component when the two RF drive signals have some imbalance. The entire output stability of the multi-carrier generator has been analyzed with different dominant crosstalk component. The experimental results are in good agreement with the theoretical results. The theoretical and experimental results may provide a useful guide for achieving the high-quality of the SSB-based RFS multi-carrier source for Tb/s multi-carrier transmission in practice.
Acknowledgements
This work is supported by National High Technology Research and Development Program of China (Grant No. 2009AA01Z224) and National Natural Science Foundation of China (Grant No. 60977049).
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