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We present automatic dispersion control of 1.28Tb/s optical time domain multiplexed signals. The dispersion is monitored by measuring the power of the 1.28THz tone of the RF spectrum using a photonic-chip-based radio-frequency spectrum analyzer (PC-RFSA) and the dispersion compensation is realized by means of a spectral pulse shaper, via computer-controlled feedback from the PC-RFSA.
© 2010 Optical Society of America
1. Introduction
Driven by the ever-increasing need for more bandwidth and a reduction of the power consumption in optical networks, the use of higher single-channel bit rates is being explored as a possible solution. Recently, optical time division multiplexing (OTDM) has been used to generate a single-channel data symbol rate exceeding 1 Terabaud [1, 2]. At such high bit rates, accurate monitoring of impairments in the signal becomes increasingly difficult. The signal characterization is generally achieved by ultra-fast nonlinear optics, most commonly for temporal waveform measurements by techniques such as intensity autocorrelation [3] or optical sampling [4,5]. Alternatively, the radio-frequency (RF) spectrum, by definition the power spectrum of the temporal intensity, can be used to monitor various impairments of optical signals [6], including dispersion, polarization-mode dispersion and timing jitter. Recently, we reported an all-optical photonic-chip-based RF spectrum analyzer which is based on cross-phase modulation in a highly nonlinear dispersion-engineered chalcogenide planar waveguide, offering THz measurement bandwidth due to reduced walk-off [7]. Among other impairments, this PC-RFSA allows the monitoring of dispersion-induced pulse broadening [8], which becomes increasingly important as bit rates of optical links keep increasing, since a fourfold increase of the bit rate corresponds to a 16-times reduction of the chromatic dispersion tolerance window [9].
Given that the dispersion in a 1-km link can vary more than 0.1ps/nm over a 60°C temperature range [10], it is clear that the variation of fiber dispersion with temperature is a vital parameter for ultrahigh bit rate systems. These systems therefore require tunable dispersion compensation elements [11–14] and an automatic dispersion compensation scheme was demonstrated in [15] for 160Gb/s transmission, by monitoring changes in residual dispersion and providing a control signal to a dispersion compensator based on a chirped fiber Bragg grating.
In this paper, we propose the use of our PC-RFSA to monitor the residual chromatic dispersion of a 1.28Tb/s OTDM signal and use the result of this measurement to perform automatic dispersion compensation by means of a Spectral Pulse Shaper (SPS) [16] as a tunable dispersion compensator. The principle is illustrated in Fig. 1, where an SPS is also used at the tranmitter side to emulate random dispersion variations in the transmission and for our proof-of-concept, the transmission link consists of 50m of standard single-mode fiber (SMF) and 10m of dispersion compensated fiber (DCF). The SPS is based on liquid crystal on silicon (LCoS) technology allowing for reconfigurable dispersion trimming using spectral phase modulation [16]. To generate or compensate dispersion, we apply a parabolically varying spectral phase which corresponds to a linearly varying group delay. The reconfiguration time of the SPS is less than 100ms and its bandwidth × dispersion product is 40ps [16]. In contrast to [2], where we used the PC-RFSA for only manual trial-and-error optimization of the transmitter, in this work we we perform variable dispersion emulation of a transmission link and we show automated PC-RFSA feedback control with a computer. Furthermore, we use a photo-detector (PD) instead of an optical spectrum analyzer (OSA), which greatly improves the potential for photonic integration of the complete monitoring tool onto a single device.
Fig. 1 Schematic principle of automatic dispersion compensation by monitoring the signal’s RF spectrum and computer-controlled feedback to a tunable dispersion compensator.
2. Generation and optimization of the 1.28Tb/s OTDM signal
The 1.28Tb/s signal is generated by using OTDM, as shown in Fig. 2. We use an active mode-locked fiber laser (MLFL) emitting ∼1.4-ps pulses at a 40-GHz repetition rate and centered at 1550nm. The pulses are subsequently compressed to 275fs using highly nonlinear fiber (HNLF) [17] after two stages. The signal power is controlled using variable optical attenuators (VOAs). An external electro-optic Mach-Zehnder modulator (MZM) is used to encode data on the pulses at 40Gb/s with a 231 – 1 pseudo-random bit sequence (PRBS). After amplification in a low-noise erbium-doped fiber amplifier (EDFA), a five-stage fiber interferometer circuit of 27 – 1 bit delay-length optically multiplexes (MUX) the pulses up to 1.28Tb/s (return-to-zero on-off keying with ∼35% duty cyle).
To optimize the equalization and synchronization of the MUX stages, we make use of our PC-RFSA as shown in the experimental setup in Fig. 3. When the signal under test is co-propagated with a continuous wave (CW) probe, new frequencies (i.e. the RF spectrum of the signal) will be generated around the CW probe due to cross-phase modulation [9] in a 7cm-long dispersion-engineered highly nonlinear chalcogenide (As2S3) planar waveguide [18]. The fabrication of the device is described in [19] and the 2μm-wide waveguide has a mode area Aeff = 1.23μm2, corresponding to a nonlinear coefficient γ = 104/W/km and an anomalous dispersion of about +29-ps/nm/km for the TM mode at 1550-nm. The polarization of the signal and the CW probe is aligned to the TM-mode of the waveguide by means of a polarization controller (PC), and the output of the waveguide is connected to an OSA. The signal and CW probe are combined by means of a wavelength division multiplexing (WDM) fiber coupler, and coupled into the waveguide using lensed fibers. The total (fiber-to-fiber) insertion loss of the waveguide is 12.7dB and the average power launched into the chip is 21dBm and 18dBm for signal and CW probe respectively.
Fig. 3 Experimental setup for photonic-chip-based RF spectrum monitoring for signal optimization and dispersion monitoring.
Figure 4 (top left graph) shows the optical spectrum of the optimized 1.28Tb/s OTDM signal, together with the CW probe at 1585nm and the XPM frequencies generated around this probe. The bottom graphs show the RF spectrum of the signal before (left) and after (right) optimization, the latter clearly showing that the MUX stages were very well aligned and that dispersion was optimized since the resulting RF spectrum is very clean with only one peak visible, i.e. the 1.28THz tone. The autocorrelation traces of the optimized 1.28Tb/s signal and of the 275fs compressed pulse are shown in the top right graph of Fig. 4.
Fig. 4 Opical spectrum (top left), autocorrelation trace (top right) and RF spectrum before (bottom left) and after (bottom right) optimization of the 1.28Tb/s signal.
3. Automatic dispersion compensation
To achieve automatic group velocity dispersion (GVD) compensation, we use a standard PD to measure the power in the 1.28THz tone (see Fig. 3). To this end, we apply a 1.7nm bandpass filter (BPF) centered around the 1.28THz tone (1596nm). Using a BPF and a PD rather than an OSA allows us to significantly speed up the monitoring time and offers a much higher potential for photonic integration into a single device. When the dispersion occuring in the link is changed (emulated by means of the SPS at the transmitter side), the measured power in the 1.28THz tone will decrease. This is illustrated in Fig. 5. If the power drops below a certain threshold value, the automatic dispersion compensation is activated. The entire process flow is shown in Fig. 6. For our proof-of-concept, we chose our threshold at 0.7dB below the optimum power. This should be validated by performing bit error rate (BER) measurements: the chosen threshold should correspond to a residual dispersion that allows error-free operation (preferably with a safety margin). The initial stepsize in our automatic dispersion compensation algorithm is chosen to be 0.01ps/nm. This stepsize is iteratively changed with a factor according to the difference in power measured before and after adjusting the dispersion (see Fig. 6).
The automatic dispersion compensating action versus time is shown in Fig. 7. This graph shows that every time we apply a dispersion change at the transmitter side, the system successfully recovers by applying the appropriate opposite dispersion at the receiver side. We applied discrete steps in the dispersion to be able to determine the response time of the system, whereas in practice the dispersion will change in a much smoother fashion since it will be induced by temperature fluctuations. The largest step in dispersion (ΔD = 0.09ps/nm) we applied corresponds to 0.8×LD in terms of dispersion length. In principle, the response time of the system is primarily determined by the reconfiguration time of the SPS and the time needed to make a power measurement, i.e. in the order of a few 100ms. In the current stage, however, the required spectral phase modulation image to be applied to the LCoS is generated at each new step and therefore requires a few seconds. This could be improved by fetching the spectral phase image from a pre-defined library stored locally. The necessary control electronics could also be integrated in the tunable dispersion compensator, eliminating the need for a separate computer.
Fig. 7 Automatic dispersion compensation result. Applied dispersion (dashed line) and measured power in the 1.28THz tone (solid line) versus time.
4. Conclusion
In conclusion, we have successfully demonstrated a flexible scheme for automatic compensation of residual dispersion in a 1.28Tb/s transmission link. Automatic GVD compensation can be achieved by monitoring the power of the 1.28THz tone of the signal’s RF spectrum using a highly nonlinear planar As2S3 waveguide, a BPF and a simple PD. This scheme also works with phase-encoded signals [20]. When using an OSA at the receiver side, other impairments such as higher-order dispersion could be monitored and compensated using an SPS, at the expense of a slower system response time. This will be the subject of further research.
Acknowledgments
This work was supported by an Australian Research Council (ARC) under the ARC Centres of Excellences Program and by a Linkage grant with Finisar Australia. The work of J. Van Erps was supported by the Fund for Scientific Research (FWO Vlaanderen) under a post-doctoral research fellowship, and additionally supported by Belspo-IAP, IWT-SBO, GOA, and the OZR of the Vrije Universiteit Brussel.
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