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Abstract
Electronically reconfigurable transmitarray (ERTA) combines the advantages of optic theory and coding metasurface mechanism with the characteristic of low-loss spatial feed and real-time beam manipulation. Designing a dual-band ERTA is challenging due to multiple factors, including large mutual coupling generated by dual-band operation and separate phase control in each band. In this paper, a dual-band ERTA is demonstrated with the capability of fully independent beam manipulation in two divided bands. This dual-band ERTA is constructed by two kinds of orthogonally polarized reconfigurable elements which share the aperture in an interleaved way. The low coupling is achieved by utilizing polarization isolation and a backed cavity connected to the ground. To separately control the 1-bit phase in each band, a hierarchical bias method is elaborately presented. As proof of concept, a dual-band ERTA prototype composed of 15 × 15 upper-band elements and 16 × 16 lower-band elements is designed, fabricated, and measured. Experimental results verify that fully independent beam manipulation with orthogonal polarization is implemented in 8.2-8.8 GHz and 11.1-11.4 GHz. The proposed dual-band ERTA may be a suitable candidate for space-based synthetic aperture radar imaging.
© 2023 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
Electronically reconfigurable transmitarray (ERTA) combines the advantages of optic theory and coding metasurface mechanism with the characteristic of low-loss spatial feed, real-time beam manipulation, and simple structure. As a low-profile, low-mass, and flat microwave lens, the ERTA can meet the application requirements of a real-time imaging system [1], vortex wave generator [2,3], and fast beam scanning [4,5]. Owing to the simple structure, the ERTA can be easily extended to the terahertz, optical, and even acoustics regimes. In the past few years, electronically steerable devices, including micro-electromechanical systems (MEMS) switches [6,7], varactors [8,9], and PIN diodes [10,11] have been proven to be a promising way to realize the ERTAs.
Based on the same operation mechanism, electronically reconfigurable reflectarray (ERRA) has been studied in many works of literature [12–15]. The metasurface-based ERRAs inherently possess metallic ground, which ensures high reflectivity while providing a shield for the bias circuit. Different from the ERRAs, the design of ERTAs is more challenging because the low-loss transmittance and the layout of the bias circuit should be considered simultaneously. Although facing greater design complexity, the ERRAs are more advantageous for practical application since the feed blockage is avoided.
The ERTA can be achieved by utilizing a receiver-transmitter metasurface [16], multi-layer frequency selective surface [17], or polarization conversion surface [18]. Thanks to a shared metallic ground between the receiving and transmitting patches, the receiver-transmitter metasurface is more convenient for designing the ERTA. With the progress in academic research and engineering application, multiple bits, high frequency, wide-angle beam scanning, and integrated multifunction have become hot spots in the ERTAs design. In order to reduce the quantization loss, 2-bit elements are demonstrated in [7,19]. Several ERTA configurations operating at Ka-band have been proposed for satellite communications [20,21]. ERTAs with wide-angle beam steering and low scan loss have also been investigated in [22,23]. An electronically reconfigurable unit cell for transmit-reflect-arrays is proposed in [24], which can independently control the reflection and transmission phase. Moreover, several prototypes of dual-polarized ERTA with the capability of polarization control and beam steering have been proposed [25–27].
Nevertheless, existing ERTAs usually operate at a single frequency band, which is hard to meet requirements in the multifunction radiator, dual-band synthetic aperture radar (SAR) [28], and multiple input multiple outputs (MIMO) systems. As a result, a dual-band ERTA is more suitable to be applied in the above-mentioned scenarios rather than an ultra-wideband (UWB) ERTA. In fact, it is hard for the ERTA to simultaneously realize 360° phase response and high transmittance in UWB operation. Furthermore, in practical applications, occupying two independent given bands is preferable to avoid additional electromagnetic wave interference.
In the open literature, a few fixed dual-band transmitarrays have been demonstrated in Ka-band [29] and Ku-band [30]. In [31] and [32], dual-band transmitarrays with beam scanning are achieved by mechanically moving the feeding source. Although mechanical scanning in a shared aperture is easy to be designed, fully independent beam manipulation is difficult to be achieved in dual bands. Till now, only an active dual-band transmitarray unit-cell based on a multi-functional SiGe BiCMOS MMIC has been assembled and measured at K/Ka-band [33]. The dual-band performance of the ERTA has not been explored due to multiple factors, including large mutual coupling generated by dual-band operation and separate phase control in each band.
In this paper, a dual-band ERTA is demonstrated with the capability of fully independent beam manipulation in two divided bands. The existing challenges of the large mutual coupling and separate phase control are well addressed. This ERTA consists of two kinds of orthogonally polarized reconfigurable elements which share the aperture in an interleaved way. Thanks to the orthogonal polarization isolation and the backed cavity of the proposed elements, the mutual coupling of two divided bands is extremely weak. Besides, a hierarchical bias method is elaborately presented to individually control PIN diodes in each band. To the authors’ best knowledge, the dual-band ERTA with fully independent beam manipulation has not been reported in the state-of-the-art. The organization of this paper is as follows. The working principle of the dual-band ERTA is shown in Section 2. The dual-band element design, simulation, and analysis are presented in Section 3. In Section 4, experimental verification of the dual-band ERTA is described. Finally, a conclusion is drawn in Section 5.
2. Working principle of dual-band ERTA
The conceptual illustration of the proposed dual-band ERTA is described in Fig. 1(a). This ERTA consists of two kinds of orthogonally polarized reconfigurable elements operating in two different bands. Here, we define the direction along the y-axis as horizontal polarization (H-pol.) and the direction along the x-axis as vertical polarization (V-pol.). The lower-band and upper-band elements operate in H-pol. and V-pol., respectively. Both the lower-band and upper-band elements possess the capability of the independent 1-bit phase regulation.
Fig. 1. The conceptual illustration of the proposed dual-band ERTA. (a) Perspective view of the dual-band ERTA, (b) the configuration of the dual-band array in an interleaved way, and (c) the lower-band, upper-band, and dual-band elements.
Two 10-dBi standard gain horns are adopted as feeding sources in the xoz plane. Two feeding sources work in H-pol. and V-pol., respectively. In order to balance the irradiation efficiency in dual-band operation, two feeding sources are shifted the same distance away from the focal point along the x-axis in opposite directions. By using two field programmable gate arrays (FPGAs), two coding channels placed in an orthogonal way are introduced to provide a bias voltage on each element independently. Since the 1-bit phase distributions of the lower-band and upper-band sub-arrays can be separately controlled by two coding channels, the independent beam manipulation is realized in two operation bands.
The lower-band sub-array, upper-band sub-array, and dual-band array are illustrated in Fig. 1(b). Based on the design concept of shared aperture, two kinds of orthogonally polarized reconfigurable elements are placed in an interleaved way. Benefiting from the interleaved arrangement in a common plane, the transmission performance of the dual-band ERTA can be simulated by using an equivalent element (denoted as dual-band element) with infinite period boundary conditions, as shown in Fig. 1(c).
3. Dual-band ERTA element
3.1 Design of dual-band ERTA element
Geometrical configurations of the lower-band and upper-band reconfigurable elements are illustrated in Fig. 2(a). The analogous structure with different dimensions achieves the lower-band and upper-band elements. The common prototype of the lower-band and upper-band elements is a reconfigurable receiver-transmitter meta-atom with a backed cavity. This structure can be regarded as two reversed patch antennas operating in receiving and transmitting modes. A metallic plane is adopted as the ground (GND) for receiving and transmitting patches. The two patches are connected by a through via in the middle of the element. The through via isolated from the metallic ground is placed for energy transfer.
Fig. 2. The proposed lower-band, upper-band, and dual-band elements. (a) Geometrical configurations of the low-band and upper-band elements, (b) a comparison between the conventional and the proposed bias structure, and (c) the layered schematic diagram of the dual-band element.
The process of transmission can be explained as follows. The receiving patch receives the electromagnetic wave from the feeding source. Then, the energy is transferred to the transmitting patch by a through via. Finally, the transmitting patch radiates the electromagnetic wave to free space. The receiving and transmitting patches can be divided into active and passive patches. The active patch is embedded with two PIN diodes (SMP1340-040LF), and the passive patch is a fixed structure. PIN 1 and PIN 2 work in opposite modes. For a positive bias voltage (denoted as state-1), PIN 1 is on, and PIN 2 is off; For a negative bias voltage (denoted as state-2), PIN 1 is off, and PIN 2 is on. 180° phase difference is achieved based on the current reverse mechanism.
To balance the structure of the ERTA, PIN diodes of the lower-band and upper-band elements are distributed in receiving and transmitting layers, respectively. Inspired by the literature [34], a backed cavity connected to the ground is designed to reduce the mutual coupling between the lower-band and upper-band elements. The periods of the lower-band and upper-band elements are 8 mm and 10 mm, respectively. Hence, the period of the dual-band element is 18 mm. Detailed geometrical parameters of the proposed lower-band and upper-band elements are exhibited in Table 1.
Table 1. Detailed parameters of the lower-band and upper-band elements
In particular, it is essential to design a bias structure for a reconfigurable transmitarray, which is required to realize independent phase control of all elements with low insertion loss. In general, several principles need to be met. 1) The bias lines are perpendicular to polarization of the radiation patch to realize polarization isolation. 2) The bias lines are narrow and close to the ground plane. 3) The bias point of the active patch should be placed where the induced current is very weak.
The traditional single-band bias structure is composed of metalized blind vias and bias lines. For a reconfigurable receiver-transmitter element, two dielectric substrate layers are usually occupied by the blind vias. One-layer blind vias are used to provide positive or negative bias voltage for driving PIN diodes, and the other-layer blind vias are used for ground connection [9,10,22,24,26], as shown in Fig. 2(b). For dual-band ERTA elements, the two-layer blind vias of dual-band bias structures with will be interlaced promiscuously, which inevitably leads to a complex fabrication process.
For the sake of simplification, an ingenious method utilizing the backed cavity of the proposed element is presented. Because the backed cavity is connected to ground, the ground connection for bias structure is easily realized by metallic patches connecting the passive patch to the backed cavity. The metallic patches are orthogonally polarized with the radiation patches, as shown in Fig. 2(a). Hence, the added metallic patch does not impact the radio frequency (RF) performance. In this way, only one-layer blind vias are required to achieve independent phase control in each frequency band, as show in Fig. 2(b).
On the other hand, the direction of the bias lines must be perpendicular to the polarization direction of controlled elements to relieve the direct current (DC) effects on RF performance. For the orthogonally polarized dual-band ERTA, the lower-band and upper-band bias lines will be perpendicular to each other. The lower-band and upper-band bias lines should be arranged in separate layers. Herein, a hierarchical bias method is elaborately presented.
The dual-band element comprises five metallic layers (M. L.) separated by three dielectric substrates (S) and two bonding layers (PP), as shown in Fig. 2(c). The top metallic layer is the receiving patches. The second metallic layer is the bias lines for the lower-band element, and the lower-band bias lines are connected to the lower-band active patch by blind vias. The top and second metallic layers are separated by a dielectric substrate (ROGERS ARLON AD255C, h = 1.524 mm, ɛr = 2.55). The third metallic layer is GND supported by a dielectric substrate (ROGERS 4350B, h = 0.1 mm, ɛr = 3.58). The GND acts as a metallic plane which provides shielding for the lower-band and upper band bias lines on both sides. In order to mitigate the impact of the bias lines on RF performance, the bias lines should be close to GND as possible. Hence, the two bonding layers (ROGERS 4450F, ɛr = 3.58) used to separate GND and bias lines are thin. The thicknesses of the upper bonding layer and lower bonding layer are 0.2 mm and 0.3 mm, respectively. The fourth metallic layer is the bias lines for the upper-band element, and the upper-band bias lines are connected to the upper-band active patch by blind vias. The fifth metallic layer is the transmitting patches. In order to maintain the symmetry of the layered structure and improve impedance match performance, the bottom dielectric substrate layer is the same as the top dielectric substrate layer. As a result, the lower-band and upper-band bias lines are perpendicular to each other and are arranged near the GND in separate layers.
3.2 Simulation and analysis of dual-band ERTA element
The dual-band element is simulated in ANSYS HFSS under infinite period boundary conditions. In order to investigate the influence of the bias lines on element transmission performance, one bias line, five bias lines, and eight bias lines were placed in the dual-band element separately. Figures 3(a) and (b) indicate that the bias lines do not deteriorate element transmission performance. The lower-band and upper-band operation bandwidths within 3-dB insertion loss are 8.2-8.7 GHz and 11.0-11.4 GHz, respectively. The upper-to-lower frequency ratio is 1.3 (11.2 GHz /8.5 GHz). As depicted in Figs. 3(c) and (d), 1-bit transmission phase resolutions are realized by switching the two operation states (S-1 or S-2) in two bands. When the state is changed for one band, the response of the other band is barely affected.
Fig. 3. The simulated transmission performance in dual bands. (a) The transmission coefficient of lower-band and dual-band elements, (b) the transmission coefficient of upper-band and dual-band elements, (c) the tunable transmission phase in the lower band, and (d) the tunable transmission phase in the upper band.
Furthermore, the individual lower-band and upper-band elements with the same periodicity of 18 mm are simulated separately. As plotted in Fig. 3(a) and (b), there are slight discrepancies between the lower-band (or upper-band) element and the dual-band element, which proves extremely weak mutual coupling between the lower and upper bands. Hence, the parameters of the lower-band and upper-band elements can be independently optimized to speed up the simulation procedure.
Figure 4 depicts the current distributions of the dual-band element, which also proves the weak mutual coupling between the lower and upper bands. The dual-band performance with orthogonal polarization is clearly observed. Under a horizontally polarized excitation at 8.5 GHz, the lower-band element possesses a strong horizontal current, while the upper-band element presents a feeble current. Under the excitation of vertical polarization at 11.2 GHz, the current is concentrated on the upper-band element and flows vertically. The weak mutual coupling is mainly due to the isolation of orthogonal polarization. Besides, the backed cavity connected to the metallic ground can further improve the isolation of the two channels.
Fig. 4. (a) The current distributions of the dual-band element under different polarized excitation at 8.5 GHz and 11.2 GHz, and (b) the equivalent model of the PIN diode.
Under the same excitation amplitude, the current flowing through the ON-state PIN diode is significantly stronger in the upper band, as shown in Fig. 4(a). The equivalent model of the PIN diode is shown in Fig. 4(b). In the ON state, the diode has parasitic resistance. As a result, the upper-band element generates a more significant insertion loss due to the ohmic loss. This analysis coordinates well with the simulation results in Fig. 3(a) and (b).
4. Dual-band ERTA design and verification
4.1 Design and fabrication of dual-band ERTA
To experimentally verify the performance in two bands, a dual-band ERTA prototype is designed, fabricated, and measured. The ERTA is composed of 16 × 16 horizontally polarized elements operating in the lower band and 15 × 15 vertically polarized elements operating in the upper band, as shown in Fig. 5(a). Two PIN diodes are integrated into each reconfigurable element. Hence, the ERTA prototype is embedded with 962 PIN diodes in total. Lower-band and upper-band element spacing are 18 mm (about 0.51λ at 8.5 GHz and 0.67λ at 11.2 GHz). The effective radiation apertures of the lower-band and upper-band sub-arrays are 280 mm × 280 mm and 260 mm × 260 mm, respectively. All elements are connected to two 16 × 16 DC bias controllers (+3 V/-3 V) by 64 8-pin connectors. In order to fix the 8-pin connectors, a 30 mm substrate is added to the periphery of the fabricated transmitarray. The overall dimension of the fabricated transmitarray is 340 mm × 340 mm.
Fig. 5. (a) The fabricated dual-band ERTA in side-view, and (b) the proposed ERTA was tested in microwave anechoic chamber.
Two identical 10-dBi horn antennas operating in X-band are adopted as feeding sources, as depicted in Fig. 5(b). Horn #1 and horn #2 operate at horizontal and vertical polarization, respectively. To be placed in the same focal plane, the feeding antennas are shifted 24 mm away from the focal point along the x-axis in opposite directions. Given the balance of irradiation efficiency and spillover efficiency, the focal length to diameter (F/D) ratios for the lower and upper bands are set to 0.6 and 0.65, respectively. As a result, the focal length (F) is 168 mm. By measuring the radiation pattern of the feeding antenna, the phase center is calculated as 8 mm away from the feeding antenna aperture [35]. Therefore, the distance between the feeding antenna aperture and the dual-band ERTA is 160 mm.
4.2 Dual-band radiation performance in the broadside
The shifted distance of the feeding source (Δx = 24 mm) should be considered when calculating the phase distributions. x (m, n) and y (m, n) denote the distance from element (m, n) to the geometric center of ERTA in two vertical directions, respectively. F (m, n) represents the distance from the phase center of the feeding source to the element (m, n), which can be calculated by (1):
The ideal compensated phase φ (m, n) can be obtained by combining formulas (2) and (3):
The 1-bit phase distributions for focused beams in the broadside at 8.5 GHz and 11.2 GHz are depicted in Fig. 6(a) and (b). Full-wave simulation of the proposed dual-band ERTA is carried out in CST Microwave Studio. Radiation patterns at 8.5 GHz and 11.2 GHz are shown in Fig. 6(c) and (d). In two operation frequencies, high-gain pencil beams with orthogonal polarization in the broadside direction are achieved.
Fig. 6. The phase distributions and radiation patterns for focused beams in the broadside. (a), (c) At 8.5 GHz; (b), (d) at 11.2 GHz.
Experimental verification of the fabricated antenna is carried out in the far-field anechoic chamber, as shown in Fig. 5(b). The simulated and measured gains in the broadside versus frequency are depicted in Fig. 7. The measured 3-dB gain bandwidths are 600 MHz (8.2-8.8 GHz) and 300 MHz (11.1-11.4 GHz), which are in good agreement with the simulation results of the element in Fig. 3. The measured gain of the horizontally polarized beam is 21.2 dBi at 8.5 GHz, and the measured gain of the vertically polarized beam is 22.7 dBi at 11.2 GHz. There is no mutual coupling between the two cross-polarized pencil beams. The dual-polarized application in a shared plane can improve the comprehensive utilization efficiency of radiation aperture.
Fig. 7. The simulated and measured broadside gain versus frequency. (a) In the lower band, and (b) in the upper band.
As depicted in Fig. 7(a), a slightly different is observed between the simulated and measured results in the lower band, which is closely related to the loss of soldering, fabrication, assembly, and measurement. Compared with the measured error in the lower band, a larger deterioration is observed in the upper band. There are two main reasons. The PIN diode is modeled as a series lumped RLC circuit in the simulation, as shown in Fig. 4(b). For the OFF state, the PIN diode is equivalent to lumped Roff = 10 Ω, L = 450 pH, and C = 0.086 pF in series. And for the ON state, the PIN diode is equivalent to lumped Ron = 1 Ω and L = 450 pH in series. These lumped circuit parameters are measured at 9.4 GHz. As a result, the variation of circuit parameters in the upper band introduces errors between the simulation and measurement.
On the other hand, the upper-band operation requires higher fabrication and measurement accuracy. Under the same fabrication and assembly conditions, the upper-band operation is more likely to produce the unexpected error. Both in two operation bands, the deviations of experimental measurement and simulation are within the tolerance of error.
4.3 Independent beam scanning performance
1-bit phase distributions of the lower-band and upper-band sub-arrays can be controlled independently. Hence, orthogonally polarized beam manipulation in two bands can be realized simultaneously. The phase distributions of the orthogonally polarized beams pointing to (135°, 30°) at 8.5 GHz and (45°, 20°) at 11.2 GHz are determined by formulas (3) and (4), as shown in Fig. 8(a) and (b). The simulated 3-D radiation patterns in the uv-plane at 8.5 GHz and 11.2 GHz are given in Fig. 8(c) and (d), respectively. Two high-gain pencil beams are realized in the predefined directions.
Fig. 8. The phase distributions and radiation patterns for the dual-band scanning beams. (a), (c) At 8.5 GHz; (b), (d) at 11.2 GHz.
The simulated and measured E-plane scanning beams are presented at two operation frequencies, as shown in Fig. 9. At 8.5 GHz, the horizontally polarized beam can scan to 50° with a scan loss of 3.6 dB. The vertically polarized beam can scan to 40°, which is limited by the large element spacing (0.67λ) at 11.2 GHz. The capability of independent beam manipulation is experimentally verified in two operation bands. Besides, by dynamically changing the coding arrangement, beam forming and vortex wave generation can be easily realized in two divided bands.
Fig. 9. Simulated and measured E-plane scanning beams at two frequencies. (a) At 8.5 GHz, and (b) at 11.2 GHz.
5. Conclusion
A dual-band ERTA with the capability of fully independent beam manipulation in two divided bands is designed, simulated, and measured in this paper. This ERTA is constructed by two kinds of orthogonally polarized reconfigurable elements which share the aperture in an interleaved way. The low coupling is achieved by utilizing polarization isolation and a backed cavity connected to the ground. A hierarchical bias method is elaborately presented to control PIN diodes in each band individually. The experimental results verify that the horizontally and vertically polarized beams can scan to 50° at 8.5 GHz and 40° at 11.2 GHz, respectively. The proposed dual-band ERTA may be a suitable candidate for space-based SAR imaging. In addition, the independent dual-band beam manipulation capability can be easily extended to uplink and downlink satellite communication in Ka/Ku-band.
Funding
Natural Science Basic Research Program of Shaanxi Province, China (20200108, 2020022, 20210110, 20220104, 2022JM-319, 2022JQ-685); National Natural Science Foundation of China (62171460).
Disclosures
The authors declare that there are no conflicts of interest related to this article.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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