Open Access
Add to BibSonomy
Abstract
In this paper, a characteristic mode rotation (CMR) method has been proposed to design a compact metasurface antenna with a low radar cross section (RCS) in a wideband. In the proposed CMR method, the incident wave dependent complex characteristic currents corresponding to the dominant characteristic modes solved by the characteristic mode method (CMM) are calculated. With the direction of the superposition of the complex characteristic currents orthogonal to that of the incident electric field in the CMR method, the metasurface subarray with wideband polarization conversion characteristic is designed. By arranging the metasurface subarray in a rotation way, a metasurface array with a compact size of 1.28λ0×1.28λ0 is designed for wideband RCS reduction. A miniature circle patch antenna is integrated with the metasurface array to achieve not only good radiation performance but also low observability for the in-band and the out-of-band of the antenna. Simulated and measured results demonstrate that the proposed miniature metasurface antenna designed by the CMR method has a good broadside radiation pattern, a maximal gain of 10.75 dB, and a -10 dB RCS reduction characteristic in the wide band of 6∼20.7 GHz with a fractional band of 110%.
© 2021 Optical Society of America under the terms of the OSA Open Access Publishing Agreement
1. Introduction
With the rapid development of stealth technology, radar cross section (RCS) reduction has caused much attentions. Low observability can be achieved by using conventional absorbing materials [1]. For example, Salisbury screen can reduce backscattering by placing a resistive sheet a quarter wavelength away from a metal surface [2], [3]. However, Salisbury screen generally operates in a narrow band and its high physical profile limits the application of Salisbury screen. In recent years, metamaterial/metasurface has demonstrated novel electromagnetic properties not found in natural materials [4–6] and therefore applied into various fields including antenna designs [7–17], absorber and cloak designs [18–23], array isolation improvement [24–29], etc. Perfect metamaterial absorber was proposed to obtain low RCS by absorbing electromagnetic waves and transforming the electromagnetic energy into heat [30–35]. An alternative approach for the RCS reduction is to redirect scattered energy away from the backscatter direction. Artificial magnetic conductors (AMC) and perfect conductor (PEC) were arranged in a checkboard configuration to reduce the RCS in a narrow band, utilizing 180° phase difference between them [36]. To broaden the bandwidth, a blended checkerboard structure consisting of two AMC structures with 180° ±30° reflection phase difference was designed for wideband RCS reduction [37,38]. Moreover, coding metamaterials with optimized arrangement of multibit units were developed to steer scattering waves in all directions, thus suppressing strong scattering lobes [39,40]. In addition, some polarization conversion metasurfaces have been designed to reduce the RCS [41–46]. When reasonably arranging metasurface elements with the polarization converting characteristic, polarization of the scattering wave is orthogonal to that of the incident wave, and meanwhile destructive interference at backscattering direction is generated. A similar implementation employing antenna array with the polarization conversion have been developed [47,48]. Although all reported designs demonstrate good RCS reduction performance, the designed structures are of large size and meanwhile the design procedures more rely on trial and error according to designers’ experience instead of physical insight to the RCS reduction.
Recently, characteristic mode method (CMM) has attracted increasing interests in various fields including antenna designs [49–52], invisibility cloak [53,54], etc., because it clearly elaborates the physical mechanism of the radiation and scattering problems. In this paper, we propose a characteristic mode rotation (CMR) method to design a metasurface structure having a compact size and wideband low RCS characteristic. The contributions of this paper have two folds: one is to propose a CMR method in terms of the complex characteristic currents obtained by the CMM and meanwhile design a metasurface array with wideband polarization conversion characteristic for the first time; the other is to develop a metasurface antenna with a compact size and broadband low RCS characteristic according to the designed polarization rotation metasurface array. In addition, the underlying physical mechanism is elaborated about the good radiation performance and low RCS characteristic at the same frequency. Good performance of the proposed design is demonstrated by the simulated and measured results.
2. Characteristic mode rotation method for a polarization conversion metasurface
2.1 Characteristic mode rotation method
The electromagnetic scattering from an object with an arbitrary shape S can be represented by an integro-differential equation and formulated as an impedance matrix according to method of moments (MoM), i.e.,
where J is a vector related to the induced surface current, and V is an excitation vector, and the impedance matrix Z isHere X and R are the imaginary and real parts of the impedance matrix, respectively.
In the CMM method, the scattering response from an arbitrary structure can be decomposed into multiple characteristic modes with characteristic currents (Jn), and their corresponding eigenvalues (λn), which satisfies the following relationship [49], [55]:
Due to the symmetry of the matrices R and X, the eigenvalues λn and the characteristic currents Jn are real. Moreover, the characteristic currents Jn are of the orthogonality property, i.e.,
Here an denotes the mode weighting coefficient (MWC) for the nth mode. With the orthogonality of the Jn, the an can be determined by the eigenvalues λn and the excitation vector V as [49,55]
It can be seen from Eq. (6) that the MWC is a complex value with the magnitude |an| and the phase φn. According to Eq. (5), the larger of the |an| is, the more efficiently its corresponding mode is scattered when the plane wave illuminates. Substituting Eq. (6) into Eq. (5), we have
Denoting the complex characteristic current ${\tilde{{\mathbf J}}_n}$ as
Equation (7) can be rewritten as
We can find from Eq. (9) that the induced surface current J can be expressed as the superposition of a group of the complex characteristic currents ${\tilde{{\mathbf J}}_n}$ multiplied by the MWC magnitudes |an|. With the J, the field scattering from the object can be [55]
2.2 Case 1: only a dominant mode
Without loss of generality, assume the mode 1 is the excited dominant mode. In this scenario, the |a1| is far larger than the magnitudes of other MWCs. According to Eq. (10) we have
In this scenario, the direction of the scattering far field Es is approximately same as that of the complex characteristic current ${\tilde{{\mathbf J}}_1}$. When the MWC phase φ1 is larger than -90° and meanwhile less than 90°, i.e., -90° ≤φ1≤90°, the direction of the ${\tilde{{\mathbf J}}_1}$ is approximately same as that of the characteristic current J1. If the phase φ1 is less than -90° or the phase φ1 is larger than 90°, i.e., φ1≤-90° or 90° ≤φ1, the direction of the ${\tilde{{\mathbf J}}_1}$ is approximately opposite to that of J1.
Especially when the direction of the incident electric field is orthogonal to that of the complex characteristic current ${\tilde{{\mathbf J}}_1}$, the direction of the resultant scattering electric field is orthogonal to that of the incident electric field. Therefore, the polarization rotation characteristic is achieved.
2.3 Case 2: multiple dominant modes
Without loss of generality, assume the first L modes are the dominant modes. It means that the MWC magnitudes of the first L modes are far larger than those of other MWCs. Therefore, the scattering field is calculated as
The direction of the Es is approximately same as that of the superposed complex characteristic currents. Similar to the above discussions, the direction of the ${\tilde{{\mathbf J}}_l}$ (l=1,…, L) can be determined by the direction of the characteristic current Jl and the MWC phase φl.
When the superposed complex characteristic current is perpendicular to the incident electric field, the resultant scattering electric field is orthogonal to the incident one, thus achieving the polarization conversion. The analysis and design for the polarization conversion characteristic based on the superposition of the complex characteristic currents is called the CMR method. It is worthwhile pointing out that the CMR method relies on the CMM method, and meanwhile is different from the CMM method. The complex characteristic currents in the CMR method involve not only the characteristic currents but also the phases of the MWCs, which can be obtained according to the CMM method.
2.4 Design of a polarization conversion metasurface
With the proposed CMR method, a metasurface subarray is designed for the polarization conversion, as shown in Fig. 1. The subarray is composed of a FR4 substrate, an air layer and a metallic ground. The metallic strip shown in Fig. 1(c) is obliquely fabricated on the bottom surface of the FR4 substrate. The CMM is performed for the metamsurface subarray, when an x-polarized plane wave is incident. The MWCs of the 30 characteristic modes are solved in a wide frequency band of 6∼22 GHz. Among the 30 modes, the MWC magnitudes of the first 5 modes are nonzero, while the MWC magnitudes of the remaining modes are close to zero. Figure 2(a) shows the MWC magnitudes of the first 5 modes, and an enlargement for the band covering 6∼7 GHz is given in Fig. 2(b). It can be observed that the MWC amplitude of each mode varies with the frequency, meaning that the dominant modes change as the frequency varies. As shown in Fig. 2(b), the MWC magnitudes of the modes 2, 3, and 4 are larger at 6 GHz, and thus we know these modes dominate. According to Fig. 2(c), we know that at 6 GHz, the MWC phases of the modes 2, 3, and 4 are -25°, -150° and -150°, respectively. Figure 3(a) shows the characteristic currents of the dominant modes. It can be seen from Fig. 3(a) that the characteristic current distribution of the mode 2 on the metallic strips of the subarray are symmetry along the diagonal line of φ=45°. The components normal to the diagonal line are cancelled out and the components parallel to the diagonal line flow along the direction of φ=225°. For the characteristic current distribution of the mode 3, the components parallel to the diagonal line flow along the direction of φ=45°, with the components normal to the diagonal line cancelled out. The characteristic current distribution of the mode 4 flows along the direction of φ=315°. Introducing the MWC phases of the modes 2, 3, and 4 into the corresponding characteristic current distributions, respectively, their complex characteristic current distributions are given in Fig. 3(b). It can be seen that the complex characteristic currents of the modes 2 and 3 flow along the direction of φ=225°, while the complex characteristic current of the mode 4 flows along the direction of φ=135°. Therefore, the superposition of three complex characteristic currents flows along –x direction. According to the above CMR method, we can know that the polarization of the scattering field is same as that of the incident field, and therefore the polarization conversion doesn’t happen. At 12 GHz, the modes 3, 4, and 5 are the dominant modes, as shown in Fig. 2(a), and the corresponding MWC phases of the dominant modes are 127°, -170°, -135°, respectively, as shown in Fig. 2(c). The characteristic current distribution of the mode 3 is symmetry and flows along the diagonal line of φ=45°, and those of the modes 4 and 5 are symmetry and flow along the diagonal line of φ=135°. Considering their MWC phases, the directions of the complex characteristic currents of the three modes are opposite to those of the corresponding characteristic currents. Therefore, the superposition of the three complex characteristic currents flows along –y direction, which achieves the polarization conversion. Following the above procedure, we can find that in a wideband of 7.3∼20.4 GHz, the direction of the superposed complex characteristic currents is perpendicular to that of the incident electric field, while out of the band, the direction of the superposed complex characteristic currents is parallel to that of the incident electric field. Figure 4 gives the polarization conversion rate (PCR) of the designed metasurface subarray. It can be seen that the PCR is larger than 0.9 in the band of 7.3∼20.4 GHz. Here the PCR denotes that the ratio of the cross polarization component of the scattering field to the total scattering field. Figure 5 gives the variation of the PCR with the geometric parameters of the unit cell. With the increases of the RL, the band for the PCR over 0.9 slightly increases. On the other hand, as the RW increases, the band for the PCR over 0.9 first increases and then decreases. The optimized RL and RW are chosen as 10.2 mm and 1.8 mm, respectively. Owing to the structure symmetry, a similar conclusion can be made for the y-polarized incoming wave.
Fig. 1. Configuration of a metasurface subarray. (a) Top view. (b) Side view. (c) Unit cell. L=32 mm, T=8 mm, RL=10.2 mm, RW=1.8 mm, SL=2.1 mm, SW=0.5 mm, SL2 = 0.4 mm, SW2 = 0.5 mm, H1 = 1 mm, H2 = 3.5 mm.
Fig. 2. MWCs of the first 5 modes of the subarray when a plane wave is incident. (a) Magnitude. (b) Enlargement for the band of 6∼7 GHz. (c) Phase.
Fig. 3. (a) Dominant characteristic current distributions of the subarray at 6 GHz. (b) Dominant complex characteristic current distributions of the subarray at 6 GHz. (c) Dominant characteristic current distributions of the subarray at 12 GHz. (d) Dominant complex characteristic current distributions of the subarray at 12 GHz.
Fig. 5. Variation of the PCR of the metasurface subarray with the geometric parameters of the unit cell.
3. Design of compact antenna with low RCS characteristic
A metasurface array with a size of 1.28λ0×1.28λ0 is generated by arranging the 2×2 metasurface subarray in a rotation way, as shown in Fig. 6(a). Here λ0 is wavelength in free space at 6 GHz. Compared with the PEC plate with the same size, the RCS of the metasurface array is decreased by -10 dB from 6.6 GHz to 20 GHz, as shown Fig. 6(b). This is because each subarray has the polarization conversion property, and meanwhile the rotation layout of the subarray results in the polarization cancellation, thus achieving its RCS reduction. Figure 7 demonstrates the effect of the geometric parameters of the metasurface array on the backscattering RCS. It can be observed that the change of the RL has an effect on the low frequency RCS, and the variation of the RW affects the high frequency RCS. When the RL=10.2 mm and RW=1.8 mm, the smallest backscattering RCS can be obtained in the entire band.
Fig. 6. (a) Configuration of the metasurface array. (b) Backscattering RCS comparison between the metasurface array and the metallic ground plate with the same size.
Fig. 7. Variation of the backscattering RCS with the geometric parameters of the metasurface array. (a) RL. (b) RW.
In order to achieve an antenna with the low RCS property, a circle microstrip antenna as the initial antenna is first designed, as shown in Fig. 8. In order to guarantee the integration of the initial antenna with the metasurface array, the initial antenna has the same substrate and ground plate as the designed metasurface array. Here the circle patch antenna with a diameter of 8 mm is fabricated on the top surface of the FR4 substrate. Three rectangular slots with a width of 0.5 mm and a length of 3 mm which are symmetrically distributed along the circumference are etched on the patch to improve the impedance matching of the antenna. A coaxial probe with a dimeter of 0.87 mm is placed at a distance of 7 mm away from the patch center to excite the antenna. The performance of the initial antenna is shown in Fig. 9. The impedance band of the initial antenna is from 6.9 GHz to 7.6 GHz, with the radiation efficiency larger than 90% and the realized gain over 8 dB.
Fig. 9. Performance of the initial microstrip antenna. (a) S11. (b) Radiation efficiency. (c) Realized gain.
Next, the initial patch antenna is integrated with the metasurface array for the design of the antenna with the low RCS. In order to reduce the effect of the metasurface array on the circular patch, some unit cells in the array are removed, as shown in Fig. 10(a). Figure 10(b) shows the RCS comparison between the metasurface antenna and the metallic ground plate with the same size. It can be found that the bandwidth of the metasurface antenna for the -10 dB RCS reduction covers 6∼20.7 GHz. Comparing Fig. 10(b) with Fig. 6(b), we can find that the frequency band of the metasurface antenna for the -10 dB RCS reduction is larger than that of the sole metasurface array. The RCS reduction better than -10 dB at 6 GHz is due to the initial patch antenna. In order to elaborate the explanation about it, the CMM analysis is implemented for the initial circle patch when a plane wave is incident on it. The 10 characteristic modes are calculated. Among these 10 modes, the MWC magnitudes of only 2 modes, i.e., the modes 1 and 3, are not close to zero. Therefore, the modes 1 and 3 are dominant scattering modes of the initial antenna. Figure 11 shows the MWC magnitudes and phases of the modes 1 and 3. At 6 GHz, the MWC phases of the modes 1 and 3 are 138° and -122°, respectively. Figure 12 shows the characteristic current distributions of the modes 1 and 3 at 6 GHz. It can be seen from Fig. 12 that the direction of the characteristic current of the modes 1 is in opposite to that of the mode 3. Considering their MWC phases and characteristic current directions, we can know that the complex characteristic currents of the modes 1 and 3 are in opposite to each other, thus giving rise to the cancellation of the resultant scattering fields at 6 GHz [38].
Fig. 10. (a) The configuration of the metasurface array based antenna. (b) Backscattering RCS comparison between the metasurface array based antenna and the metallic ground plate with the same size.
Fig. 11. The CMM analysis of the initial circular patch when a plane wave is incident. (a) The magnitudes of MWCs of the dominant modes. (b) The phases of MWCs of the dominant modes.
Fig. 12. (a) The characteristic current distribution of the mode 1. (b) The characteristic current distribution of the mode 3.
Figure 13 gives RCS comparison between the initial circle patch antenna without and with the metasurface array for the incoming waves with the different incident angles at 12 GHz. It can be observed that regardless of normal and oblique incidences, with the metasurface array, the strong backscattering fields are transformed into four weaker beams in different directions, thus giving rise to the RCS reduction.
Fig. 13. Comparison of 3D scattering patterns between the initial antenna without and with the metasurface array for different incident angles at 12 GHz. (a) The initial antenna for normal incidence. (b) The initial antenna for oblique incidence with the angle of -30°. (c) The metasurface antenna for normal incidence. (d) The metasurface antenna for oblique incidence with the angle of -30°.
4. Simulation and measurement results
The proposed metasurface antenna with the low RCS property is fabricated, as shown in Fig. 14(a), and the radiation and scattering performance are measured in the anechoic chamber, respectively, as shown in Fig. 14(b). The measured operation band of the proposed antenna for S11≤-10 dB covers 6.08∼7.44 GHz, in agreement with the simulation one of 6.72∼7.4 GHz, as shown in Fig. 15(a). The slight discrepancy between them is the fabrication errors including the substrate and the coaxial probe welding. It is worthwhile pointing out that with the use of the metasurface array, the operating band of the initial circular antenna moves towards low frequencies, thus achieving the miniaturization design. Figure 15(b) gives the measured and simulated gains of the proposed antenna. A good agreement between them can be observed. The maximal gain of 10.75 dB is achieved at 7.4 GHz. Note that the measured return loss of the metasurface antenna is slightly better than the simulated one, while the measured gain is approximately same as the simulated gain. This is because some losses in the fabrication procedure are introduced. Figure 16 shows the simulated and measured radiation patterns in YOZ plane at 6.7 GHz, 6.9 GHz, 7 GHz, and 7.3 GHz, respectively, in good agreement between each other. The maximal radiation is in the broadside direction. Figure 10(b) gives the simulated and measured backscattering RCSs of the proposed antenna. The band for -10 dB RCS reduction is 6∼20.7 GHz with a fractional band of 110%, simultaneously covering the in-band and out-of-band of the metasurface antenna.
Fig. 14. (a) Bottom and front views of the fabricated metasurface antenna. (b) Measurement environment.
Fig. 16. Radiation patterns of the proposed metasurface antenna at different frequencies in yoz plane. (a) 6.7 GHz. (b) 6.9 GHz. (c) 7 GHz. (d) 7.3 GHz.
According to above discussions about the complex characteristic currents in the scattering scenario, we can know that the RCS reduction of the metasurface antenna at 6 GHz is due to the circular patch antenna. On the other hand, the measured impedance band of the metasurface antenna is from 6.08 GHz to 7.44 GHz. It means that at 6 GHz, the circular patch antenna has good radiation performance. In order to elaborate the radiation performance, a similar CMM analysis of the initial circular patch antenna is performed. Specifically, the 10 characteristic modes are calculated in the band of 5∼8 GHz when the coaxial probe of the initial antenna is excited. Among these 10 modes, the MWC magnitudes of the 7 modes are close to zero. Figure 17(a) show the 3 nonzero MWC magnitudes, i.e., modes 2, 3, and 4. Compared with the mode 3, the modes 2 and 4 are the dominant radiation modes. Figure 17(b) demonstrates the MWC phases of the dominant radiation modes. The MWC phases of the modes 2 and 4 are 150° and -115° at 6 GHz, respectively. The characteristic current distribution of the mode 2 is approximately same as that of the mode 4 at 6 GHz, as shown in Fig. 18. The resulting radiation fields of the modes 2 and 4 are superposed at 6 GHz, thus giving rise to good radiation performance. It is worthwhile pointing out that in the scattering scenario, the modes 1 and 3 are excited, while in the radiation scenario, the modes 2 and 4 are excited. Therefore, with different excited characteristic modes, the initial antenna has not only good RCS reduction characteristic but also good radiation performance at the same frequency.
Fig. 17. The CMM analysis of the initial circular patch when a coaxial probe is excited. (a) The magnitudes of MWCs of the nonzero 3 modes. (b) The phases of MWCs of the dominant modes.
Fig. 18. (a) The characteristic current distribution of the mode 2. (b) The characteristic current distribution of the mode 4.
Table 1 gives the RCS reduction comparison between the proposed design and the reported designs with the polarization conversion characteristic. Compared with the published structures, the proposed design has not only a wide RCS reduction band but also a miniature size.
Table 1. Performance comparison between the proposed and reported designs.
5. Conclusions
In this paper, a characteristic mode rotation method has been presented to design the metasurface antenna with the low RCS property. By analyzing the complex characteristic currents, the designed metasurface subarray has a polarization conversion property in a wide band. By arranging the subarray in a rotation way and integrating the resultant array with a miniature circle patch, a metasurface antenna with the low RCS characteristic is obtained in a wide band of 6∼20.7 GHz with a fractional band of 110%, simultaneously achieving in-band and out-of-band RCS reduction. The proposed antenna has a compact size of 1.28λ0×1.28λ0, good broadside radiation pattern, and the maximal gain of 10.75 dB compared with the reported designs, which provides a promising way for the miniature antenna designs with broadband low RCS property.
Funding
Shaanxi Outstanding Youth Science Foundation; National Natural Science Foundation of China (61771359).
Disclosures
The authors declare no conflicts of interest.
References
1. K. J. Vinoy and R. M. Jha, Radar Absorbing Materials: From Theory to Design and Characterization. (Kluwer, USA, 1996)
2. W. W. Salisbury, “Absorbent body for electromagnetic waves,” U.S. Patent 2 599 944, Jun. 10, 1952.
3. R. L. Fante and M. T. McCormack, “Reflection properties of the Salisbury screen,” IEEE Trans. Antennas Propag. 36(10), 1443–1454 (1988). [CrossRef]
4. Y. Shi, T. Hao, L. Li, and C. H. Liang, “An improved NRW method to extract electromagnetic parameters of metamaterials,” Microw. Opt. Technol. Lett. 58(3), 647–652 (2016). [CrossRef]
5. Y. Shi, Z. Y. Li, L. Li, and C. H. Liang, “An electromagnetic parameters extraction method for metamaterials based on phase unwrapping technique,” Waves Random Complex Media 26(4), 417–433 (2016). [CrossRef]
6. Y. Shi, Z. Y. Li, L. Li, and C. H. Liang, “A retrieval method of effective electromagnetic parameters for inhomogeneous metamaterials,” IEEE Trans. Microw. Theory Techn. 65(4), 1160–1178 (2017). [CrossRef]
7. M. Alibakhshikenari, B. S. Virdee, S. Aissa, C. H. See, A. A. Althuwayb, R. Abd-Alhameed, I. Huynen, F. Falcone, and E. Limiti, “Singular integral formulations for electrodynamic analysis of metamaterial-inspired antenna array,” IEEE Antennas Propag. Lett. 20(2), 179–183 (2021). [CrossRef]
8. M. Alibakhshikenari, F. Babaeian, B. S. Virdee, S. Alssa, L. Azplicueta, C. H. See, A. A. Althuwayb, I. Huynen, R. Abd-Alhameed, F. Falcone, and E. Limiti, “A comprehensive survey on “Various decoupling mechanisms with focus on metamaterial and metasurface principles applicable to SAR and MIMO antenna systems,” IEEE Access 8, 192965–193004 (2020). [CrossRef]
9. M. Alibakhshikenari, S. B. Virdee, A. A. Althuwayb, S. Alssa, C. H. See, R. Abd-Alhameed, F. Falcone, and E. Limiti, “Study on on-chip antenna design based on metamaterial-inspired and substrate-integrated waveguide properties for millimetre-wave and THz integrated-circuit applications,” J. Infrared Millim. Terahz Waves 42(1), 17–28 (2021). [CrossRef]
10. M. Alibakhshikenari, B. S. Virdee, P. Shukla, N. O. Parchin, L. Azpilicueta, C. H. See, R. Abd-Alhameed, F. Falcone, I. Huynen, T. A. Denidni, and E. Limiti, “Metamaterial-inspired antenna array for application in microwave breast imaging systems for tumor detection,” IEEE Access 8, 174667–174678 (2020). [CrossRef]
11. M. Alibakhshikenari, B. S. Virdee, L. Azpilicueta, M. Naser-Moghadasi, M. O. Akinsolu, C. H. See, B. Liu, R. Abd-Alhameed, F. Falcone, I. Huynen, T. A. Denidni, and E. Limiti, “A comprehensive survey of “Metamaterial transmission-line based antennas: design, challenges, and applications,” IEEE Access 8, 144778–144808 (2020). [CrossRef]
12. M. Alibakhshikenari, B. S. Virdee, P. Shukla, C. H. See, R. Abd-Alhameed, F. Falcone, and E. Limiti, “Improved adaptive impedance matching for RF front-end systems of wireless transceivers,” Sci. Rep. 10(1), 14065 (2020). [CrossRef]
13. M. Alibakhshikenari, B. S. Virdee, M. Khalily, C. H. See, R. Abd-Alhameed, F. Falcone, T. A. Denidni, and E. Limiti, “High-gain on-chip antenna design on silicon layer with aperture excitation for terahertz applications,” IEEE Antennas Wirel. Propag. Lett. 19(9), 1576–1580 (2020). [CrossRef]
14. M. Alibakhshikenari, B. S. Virdee, C. H. See, P. Shukla, S. Salekzamankhani, R. Abd-Alhameed, F. Falcone, and E. Limiti, “Study on improvement of the performance parameters of a novel 0.41-0.47 THz on-chip antenna based on metasurface concept realized on 50µm GaAs-layer,” Sci. Rep. 10(1), 11034 (2020). [CrossRef]
15. M. Alibakhshikenari, B. S. Virdee, C. H. See, R. Abd-Alhameed, F. Falcone, and E. Limiti, “High-gain metasurface in polyimide on-chip antenna based on CRLH-TL for sub terahertz integrated circuits,” Sci. Rep. 10(1), 4298 (2020). [CrossRef]
16. Y. Shi, K. Li, J. Wang, L. Li, and C. H. Liang, “An etched planar metasurface half maxwell fish-eye lens antenna,” IEEE Trans. Antennas Propag. 63(8), 3742–3747 (2015). [CrossRef]
17. Y. Shi and Y. C. Li, “Design of broadband leaky-wave antenna based on permeability-negative transmission line,” Microw. Opt. Technol. Lett. 60(3), 699–704 (2018). [CrossRef]
18. Y. Shi and Y. Zhang, “Design of Graphene-based Metamaterial Absorber and Antenna,” in Metamaterials and Metasurface, edited by J. Canet-Ferrer, (IntechOpen, United Kingdom, 2019, 171–912)
19. L. Zhang, Y. Shi, and C. H. Liang, “Achieving illusion and invisibility of inhomogeneous cylinders and spheres,” J. Opt. 18(8), 085101 (2016). [CrossRef]
20. L. Zhang, Y. Shi, and C. H. Liang, “Optimal illusion and invisibility of multilayered anisotropic cylinders and spheres,” Opt. Express 24(20), 23333–23352 (2016). [CrossRef]
21. Y. Shi, W. Tang, L. Li, and C. H. Liang, “Three-dimensional complementary invisibility cloak with arbitrary shapes,” IEEE Antennas Wirel. Propag. Lett. 14, 1550–1553 (2015). [CrossRef]
22. Y. Shi, L. Zhang, W. Tang, L. Li, and C. H. Liang, “Design of a minimized complementary illusion cloak with arbitrary position,” Int. J. Antennas Propag. 2015, 1–7 (2015). [CrossRef]
23. Y. Shi, W. Tang, and C. H. Liang, “A minimized invisibility complementary cloak with a composite shape,” IEEE Antennas Wirel. Propag. Lett. 13, 1800–1803 (2014). [CrossRef]
24. M. Alibakhshikenari, B. S. Virdee, and E. Limiti, “Study on isolation and radiation behaviours of a 34×34 array-antennas based on SIW and metasurface properties for applications in terahertz band over 125-300 GHz,” Optik 206, 163222 (2020). [CrossRef]
25. M. Alibakhshikenari, B. S. Virdee, P. Shukla, C. H. See, R. Abd-Alhameed, F. Falcone, K. Quazzane, and E. Limiti, “Isolation enhancement of densely packed array antennas with periodic MTM-photonic bandgap for SAR and MIMO Systems,” IET Microw. Antenna Propag. 14(3), 183–188 (2020). [CrossRef]
26. M. Alibakhshikenari, A. Salvucci, G. Polli, B. S. Virdee, C. H. See, R. Abd-Alhameed, F. Falcone, A. ANdujar, and J. Anguera, E. Limiti, “Mutual coupling reduction using metamaterial supersubstrate for high performance & densely packed planar phased arrays,” 2018 22nd International Microwave and Radar Conference, 675–678 (2018).
27. M. Alibakhshikenari, B. S. Virdee, C. H. See, R. Abd-Alhameed, F. Falcone, and E. Limiti, “Array antenna for synthetic aperture radar operating in X and Ku-bands: a study to enhance isolation between radiation elements,” 12th European Conference on Synthetic Aperture Radar, 1083–1087 (2018).
28. M. Alibakhshikenari, B. S. Virdee, C. H. See, R. Abd-Alhameed, F. Falcone, and E. Limiti, “Surface wave reduction in antenna arrays using metasurface inclusion for MIMO and SAR systems,” Radio Sci. 54(11), 1067–1075 (2019). [CrossRef]
29. M. Alibakhshikenari, B. S. Virdee, C. H. See, R. Abd-Alhameed, F. Falcone, and E. Limiti, “High-isolation leaky-wave array antenna based on CRLH metamaterial implemented on SIW with ±30° frequency beam-scanning capability at millimeter-waves,” Electron. 8(6), 642 (2019). [CrossRef]
30. N. I. Landy, S. Sajuyigbe, J. J. Mock, D. R. Smith, and W. J. Padilla, “Perfect metamaterial absorber,” Phys. Rev. Lett. 100(20), 207402 (2008). [CrossRef]
31. L. Li, Y. Yang, and C. H. Liang, “A wide-angle polarization-insensitive ultra-thin metamaterial absorber with three resonant modes,” J. Appl. Phys. 110(6), 063702 (2011). [CrossRef]
32. Y. Shi, J. Yang, H. Shen, Z. K. Meng, and T. Hao, “Design of broadband metamaterial-based ferromagnetic absorber,” Mater. Sci. Adv. Composite Mater. 2, 1–7 (2018). [CrossRef]
33. Y. Zhang, Y. Shi, and C. H. Liang, “A broadband tunable graphene-based metamaterial absorber,” Opt. Mater. Express 6(9), 3036–3044 (2016). [CrossRef]
34. Y. Shi, Y. C. Li, T. Hao, L. Li, and C. H. Liang, “A design of ultra-broadband metamaterial absorber,” Waves Random Complex 27(2), 381–391 (2017). [CrossRef]
35. L. Zhang, Y. Shi, J. X. Yang, X. Zhang, and L. Li, “Broadband transparent absorber based on indium tin oxide-polyethylene terephthalate film,” IEEE Access 7, 137848–137855 (2019). [CrossRef]
36. M. Paquay, J. C. Iriarte, and I. Ederra, “Thin AMC structure for radar cross-section reduction,” IEEE Trans. Antennas Propag. 55(12), 3630–3638 (2007). [CrossRef]
37. W. Chen, C. A. Balanis, and C. R. Birtcher, “Checkerboard EBG surfaces for wideband radar cross section reduction,” IEEE Trans. Antennas Propag. 63(6), 2636–2645 (2015). [CrossRef]
38. Y. Shi, Z. K. Meng, W. Y. Wei, W. Zheng, and L. Li, “Characteristic mode cancellation method and its application for antenna RCS reduction,” IEEE Antennas Wireless Propag. Lett. 18(9), 1784–1788 (2019). [CrossRef]
39. H. Zhang, Y. Lu, J. Su, Z. Li, J. Liu, and Y. Yang, “Coding diffusion metasurfaces for ultra-wideband RCS reduction,” Electron. Lett. 53(3), 187–189 (2017). [CrossRef]
40. J. F. Han, X. Y. Cao, J. Gao, J. Wei, Y. Zhao, S. Li, and Z. Zhang, “Broadband radar cross section reduction using dual-circular polarization diffusion metasurface,” IEEE Antennas Wireless Propag. Lett. 17(6), 969–973 (2018). [CrossRef]
41. Y. Jia, Y. Liu, Y. J. Guo, K. Li, and S. X. Gong, “A dual-patch polarization rotation reflective surface and its application to ultra-wideband RCS reduction,” IEEE Trans. Antennas Propag. 65(6), 3291–3295 (2017). [CrossRef]
42. R. Zaker and A. Sadeghzadeh, “A low-profile design of polarization rotation reflective surface for wideband RCS reduction,” IEEE Antennas Wireless Propag. Lett. 18(9), 1794–1798 (2019). [CrossRef]
43. Y. Lu, J. X. Su, J. B. Liu, Q. X. Guo, H. C. Yin, Z. R. Li, and J. M. Song, “Ultrawideband monostatic and bistatic RCS reductions for both copolarization and cross polarization based on polarization conversion and destructive interference,” IEEE Trans. Antennas Propag. 67(7), 4936–4941 (2019). [CrossRef]
44. Y. Liu, K. Li, Y. T. Jia, Y. W. Hao, and S. X. Gong, “Wideband RCS reduction of a slot array antenna using polarization conversion metasurfaces,” IEEE Trans. Antennas Propag. 64(1), 326–331 (2016). [CrossRef]
45. S. Y. Sun, W. Jiang, X. Q. Li, P. Liu, and S. X. Gong, “Ultrawideband high-efficiency 2.5-dimensional polarization conversion metasurface and its application in RCS reduction of antenna,” IEEE Antennas Wireless Propag. Lett. 18(5), 881–885 (2019). [CrossRef]
46. T. Hao, S. Wang, Z. Y. Li, and S. X. Gong, “RCS reduction and gain enhancement for the circularly polarizaed array by polarization conversion metasurface coating,” IEEE Antennas Wireless Propag. Lett. 18(1), 167–171 (2019). [CrossRef]
47. Y. Shi, X. F. Zhang, Z. K. Meng, and L. Li, “Design of low-RCS antenna using antenna array,” IEEE Trans. Antennas Propag. 67(10), 6484–6493 (2019). [CrossRef]
48. Z. K. Meng, Y. Shi, W. Y. Wei, and X. F. Zhang, “Multifunctional scattering antenna array design for orbital angular momentum vortex waves and RCS reduction,” IEEE Access 8, 109289–109296 (2020). [CrossRef]
49. M. Cabedo-Fabres, E. Antonino-Daviu, A. Valero-Nogueira, and M. F. Batalle, “The theory of characteristic modes revisited: A contribution to the design of antennas for modern applications,” IEEE Antennas Propag. Mag. 49(5), 52–68 (2007). [CrossRef]
50. K. Li and Y. Shi, “Wideband MIMO handset antenna design based on theory of characteristic modes,” Int. J. RF Microw. Comput. Aid Eng. 28(4), e21217 (2018). [CrossRef]
51. K. Li and Y. Shi, “A pattern reconfigurable MIMO antenna design using characteristic modes,” IEEE Access 6, 43526–43534 (2018). [CrossRef]
52. K. Li, Y. Shi, H. Shen, and L. Li, “A characteristic-mode-based polarization-reconfigurable antenna and its array,” IEEE Access 6, 64587–64595 (2018). [CrossRef]
53. L. Zhang and Y. Shi, “Bifunctional arbitrarily-shaped cloak for thermal and electric manipulations,” Opt. Mater. Express 8(9), 2600–2613 (2018). [CrossRef]
54. Y. Shi and L. Zhang, “Cloaking design for arbitrarily shape objects based on characteristic mode method,” Opt. Express 25(26), 32263–32279 (2017). [CrossRef]
55. R. F. Harrington and J. R. Mautz, “Theory of characteristic modes for conducting bodies,” IEEE Trans. Antenn. Propag. 19(5), 622–628 (1971). [CrossRef]
