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We present a reach-extender for the upstream transmission path of 10Gb/s passive optical networks based on an optimised cascade of two semiconductor optical amplifiers (SOAs). Through careful optimisation of the bias current of the second stage SOA, over 19dB input dynamic range and up to 12dB compression of the output dynamic range were achieved without any dynamic control. A reach of 70km and split up to 32 were demonstrated experimentally using an ac-coupled, continuous-mode receiver with a reduced 56ns ac-coupling constant.
©2011 Optical Society of America
1. Introduction
Today, there is high interest in new generations of passive optical networks (PONs) with 10Gb/s line-rates for the downstream and upstream directions. For long reach applications [1], this requires the development of mid-span reach-extenders to increase the PON optical budget to support the additional insertion loss of the trunk fibres. Semiconductor optical amplifiers (SOAs) have been shown to be good candidates for reach-extenders in conventional Gigabit PONs [2] and several authors have considered their use in 10Gb/s PONs [3–5]. A theme in [3–5] is to also use the SOA to compress the input dynamic range of the upstream channel (which results from differential access loss and variations in the launched power from the optical network unit (ONU)), thus reducing the required dynamic range for the burst-mode receiver (BMRx) located in the optical line termination (OLT). In [3], it is proposed to boost the signal and compress the input dynamic range by operating the SOAs in deep saturation for strong input signals. However, with conventional SOAs, operation in deep saturation leads to large overshoots on rising edges in the input signal and eye closure due to gain compression and recovery effects (which have a time constant of a few hundred picoseconds, comparable to the bit period at 10Gb/s). The authors in [3] therefore propose to use an SOA with an increased nanosecond gain recovery time and show theoretically that this reduces the signal quality degradation. However, standard SOAs do not exhibit such long gain recovery times. Another solution uses a narrow optical filter at the receiver side to reduce the signal distortion by suppressing the broadening caused by the chirped components of the signal in the frequency domain [4]. However, this introduces loss and would require the use of tightly wavelength-specified, temperature-controlled lasers in the ONU, which are expensive. In contrast, recent 10Gb/s PON standards specify a wide 20nm wavelength band for the upstream channel to allow the use of uncooled lasers at the ONU. In [5], it is proposed to switch the SOA bias current based on the magnitude of the incoming packets. While this allows significant compression of the input dynamic range while avoiding saturation of the SOA, it requires a monitoring photodiode and high-speed electronics to adjust the bias current on a packet basis.
This paper demonstrates the feasibility of an upstream reach-extender based on an optimised cascade of two SOAs that overcomes the disadvantages of the above methods. The scheme gives sufficient dynamic range compression without any dynamic control to enable the use of an ac-coupled, continuous-mode receiver at the optical line termination.
2. Principle of operation, experimental setup and results
Figure 1 shows the experimental setup. The ONU employed a 1310nm DFB laser (as a 1270nm laser was not available for this experiment) which was directly modulated at a line-rate of 10.3125Gb/s (NRZ, 231-1 PRBS, 6.5dB extinction ratio, close to the worst-case 10GEPON specification). An SOA whose bias current was modulated on a packet basis was used to emulate an alternating sequence of loud (corresponding to the maximum expected input power to the reach-extender) and soft (corresponding to the minimum expected input power to the reach-extender) packets thus emulating two ONUs with different path losses to the OLT. For correct emulation, a 1nm wide optical filter was used to suppress the ASE noise from this gated SOA. A variable optical attenuator was used to emulate splitting loss in the PON distribution network. The reach-extender was located between the distribution network (split: 32, reach: 10km) and a 60km trunk fibre. Due to differential access loss, the power of packets at the input of the reach-extender was assumed to range from −25dBm (soft packets) to −6dBm (loud packets) (IEEE 802.3av class PR20 [6]). The reach-extender consisted of a cascade of two SOAs. The first SOA provided low-noise amplification; the second SOA provided additional compression of the input dynamic range and boosted the signal launched into the trunk fibre. The OLT comprised an SOA preamplifier, a 17nm bandpass filter for out-of-band amplified spontaneous emission (ASE) suppression (a 20nm filter was not available for this experiment, however this does not alter the outcomes of this experiment), and a conventional continuous-mode PIN receiver with a reduced 56ns ac-coupling time constant (560pF coupling capacitor) to make it suitable for burst-mode operation with less than 800ns preamble (10GEPON standard). A variable optical attenuator was added in front of the PIN receiver to measure power penalties. The SOAs used in the experiments had + 21dB small signal gain, −3dB gain saturation output power of + 11dBm, 1dB polarisation dependent gain (PDG) and a 7dB noise figure at 1310nm wavelength and 250mA bias. The polarisation controllers were for experimental characterisation purposes only and would not be required in a real deployed system, where the reach-extender would need to provide sufficient system margin for all input polarisation states, including the worst-case state.
Fig. 1 Experimental setup (DML = Directly Modulated Laser, PG = Pattern Generator, PC = Polarisation Controller, VOA = Variable Optical Attenuator, RX = Receiver, CR = Clock Recovery, LA = Limiting Amplifier, ED = Error Detector, DR = Dynamic Range).
The reach-extender was first characterised in continuous-mode and in a back-to-back (B2B) configuration without the fibres and the OLT preamplifier SOA. A mid-stage optical filter, which is typically used to remove out-of-band ASE noise from the first SOA, was intentionally omitted here. Note that in order to satisfy the specifications in [6] for a 10GEPON system (which defines the upstream wavelength window from 1260 to 1280nm), the bandwidth of any mid-stage filter needs to be at least 20nm wide. As is evident from the eye diagrams shown in Fig. 2a (measured at the output of the reach-extender for soft packets, bias current of the 2nd stage SOA: 250mA), significant reduction of the eye-closure induced by patterning can be achieved by omitting such a filter. Patterning occurs when the input signals at a bit rate comparable to the gain recovery time are sufficiently strong to saturate the SOA, leading to carrier depletion in the gain medium and consequently reducing the optical gain. The signal-dependent gain of the saturated SOA leads to distortions at the bit level, which results in a signal-dependent extinction ratio degradation and intersymbol interference (ISI) [7,8]. The observed reduction in patterning can be explained by noting that for soft packets the input to the 2nd stage SOA mainly consists of ASE noise (−4.5dBm signal power, +2.7dBm ASE power). The total power is sufficiently high to saturate the 2nd stage SOA. The ASE noise acts as a continuous-wave (CW) ‘holding’ beam, which is known to speed-up the gain recovery in the SOA and reduces the patterning [9]. No similar improvement was observed for loud packets, which is consistent with the above explanation as the input to the 2nd stage SOA is then dominated by the signal rather than the ASE from the first SOA. Hence, a mid-stage filter is unnecessary for loud packets and its omission is shown to enhance the performance in the soft packets case, besides being a more cost-effective solution.
Fig. 2 (a): Soft packet eye diagrams at the reach-extender output (back-to-back), (b): gain of the 2nd stage SOA vs. input power (solid line) and Pinsat for different bias currents (dots), (c): input saturation power of the 2nd stage SOA vs. bias current.
In principle, SOA gain saturation can be used to reduce the dynamic range that must be supported by the OLT receiver if the concomitant patterning effect can be ameliorated. We demonstrate here that this is achievable by reduction of the bias current of the 2nd stage SOA. This allows significant reduction of patterning induced penalties stemming from deep saturation of the 2nd stage SOA, while at the same time maintaining essentially the same amount of input dynamic range compression. An explanation for this is provided hereafter. The large-signal gain of an SOA can be written as [10]:
where G represents the amplifier gain, G0 its unsaturated value, Pin the input power of the signal being amplified, and Psat the saturation power which depends on the gain-medium properties. The input saturation power Pinsat of an amplifier can be defined as the input power for which the gain G is reduced by half (or by 3dB) from its unsaturated value G0, and can be derived from Eq. (1) by using G = G0 / 2:where it is indicated explicitly that G0 and Psat are both dependant on the amplifier bias settings Ibias. Both Eq. (1) and (2) agree with the experimental data shown respectively in Fig. 2b and 2c, where G0 and Psat were measured for different values of the 2nd stage SOA bias current (15, 17.5, 20, 25, 30, 40, 50, 100, 150, 250mA). Figure 2b shows the well-known relation between the gain of the amplifier and its input power. The input saturation powers Pinsat are emphasised with a circular marker for each bias current. From this set of curves one can see that Pinsat rapidly decreases with increasing SOA bias current. This is especially clear when plotting Pinsat as a function of bias current as shown in Fig. 2c. For the maximum bias current of 250mA Pinsat is −6dBm, while for a bias current of 15mA Pinsat increases up to + 4dBm. In other words, if the bias current is reduced from 250 to 15mA, the 2nd stage SOA is able to tolerate a 10dB stronger input signal before entering the saturation regime. Reducing the bias current of the 2nd stage SOA may degrade the overall noise figure, however here such degradation is negligible. Indeed the noise figure NFtot of the cascaded SOA equals to [11]:where NF1 and G1 are respectively the noise figure and the gain of the first SOA, and NF2 the noise figure of the second SOA. The approximation holds when the gain of the first SOA is sufficiently large and the noise figure of the 2nd SOA reasonably small (which is the case here as the gain of the 1st SOA is greater than 20dB, and the noise figure of the 2nd SOA equals 7dB). In the worst-case a negligible 0.25dB degradation of the overall noise figure was measured for a reduction of the bias current from 250mA to 15mA.Note that reduction of the bias current reduces the power launched into the trunk fibre, but this is acceptable as long as the input power to the OLT receiver remains higher than its sensitivity. Obviously, this requirement is most stringent for the soft packet. With the setup shown in Fig. 1, in order to support 60km trunk fibre for the soft packet, it was found that a launched power at the output of the cascade of at least 0dBm is sufficient (see below). Figure 3a shows the output power of the reach-extender for both loud and soft packets. For the loud packets sufficient launch power is always available, however for the soft packets the bias current of the 2nd stage SOA needs to be at least 30mA. Figure 3a also shows the output dynamic range at the reach-extender output for a 19dB input dynamic range. It can be seen how larger dynamic range compression is achieved for higher bias currents. For bias currents higher than 30mA an almost constant 12dB (~19dB-7dB) compression of the input dynamic range can be observed.
Fig. 3 Reach-extender characteristics in B2B, (a): dynamic range and output power for soft packets (SP) and loud packets (LP), (b): soft packets and loud packets power penalties.
Next we obtained the power penalty induced by patterning by measuring the power on the PIN photodiode (without SOA3) required to achieve a bit-error rate (BER) of 1.1 x 10−3 (FEC threshold for RS(255,223) encoding, as specified in 10GEPON), and comparing it to the receiver sensitivity. The results, which were taken in continuous-mode and B2B for both soft and loud packet power levels, are shown in Fig. 3b plotted as a function of the 2nd stage SOA bias current. Up to 5dB reduction in patterning-penalty can be obtained for the loud packets by reducing the 2nd stage SOA bias current from 250 to 30mA. From Fig. 3a, this results in a negligible 0.3dB reduction in output dynamic range compression. The soft packet output power is reduced to + 1dBm at 30mA bias, which results in an OLT input power of −20dBm if a 60km trunk fibre with 21dB (0.35dB/km) insertion loss is used. To avoid additional bit errors due to thermal noise of the receiver electronics, this requires the use of an avalanche photodiode (APD) receiver to ensure operation sufficiently above the receiver sensitivity. As a suitable ac-coupled APD receiver was not available for this experiment, an SOA preamplifier (SOA3) was used instead.
Next we used the optimised cascade in a burst-mode transmission experiment, designed to emulate a 10GEPON system. Sufficient reduction in dynamic range was obtained to enable the use of a conventional ac-coupled continuous-mode PIN receiver (albeit with a reduced 56ns ac-coupling constant) followed by a conventional 10Gb/s limiting amplifier which equalises the different amplitudes of the packets after detection. When operating ac-coupled receivers in burst-mode, the ac-coupling time constant sets the required time needed to discharge the coupling capacitor after the end of a burst, and charge the coupling capacitor for a new burst. The worst-case situation occurs when a soft burst occurs immediately after a loud burst. It can be shown that the recovery time for a given power penalty αR (expressed as a linear value) is then given by [12]:
where τ is the RC time constant of the ac-coupling network, ER is the extinction ratio (defined as the power in a 1 to the power in a 0) and β is the is the loud/soft ratio (expressed as a linear value). For a negligible power penalty of 0.5dB, an extinction ratio of 6.5dB, a dynamic range of 7.4dB at the input of the receiver and the mentioned 56ns ac-coupling constant, the recovery time is calculated to be 292ns which is well in line with the 10GEPON standard (which limits the preamble for recovery of the receiver to 800ns). Note that if the ac-coupled receiver is required to handle the entire dynamic range of 19dB, the recovery time would be 452ns, hence the achieved dynamic range compression allows a significant reduction in the required receiver recovery time. It is known that a smaller ac-coupling capacitor increases the baseline wander and incurs a power penalty. This power penalty αP can be calculated as [12]:where CBDAV is the averaged cumulative bit difference in the received bit sequence (difference between number of transmitted 1s and number of transmitted 0s) [13] and Tb the bit period. For the 64B/66B scrambling specified in the 10GEPON standard, CBDAV is 8, resulting in a negligible 0.1dB penalty due to baseline wander. Figure 4a shows the structure of the packet signal used to evaluate the performance of the network at a bit rate of 10.3125Gb/s. Each packet (2040ns) consists of a preamble (800ns, alternating ‘1010’ pattern) followed by the BM synchronisation pattern, a burst delimiter and a data payload. The data payload (1240ns) consists of 231-1 PRBS sequences, additionally stressed with 66 consecutive 1s and 66 consecutive 0s. The pattern terminates with an end burst delimiter. The guard time between successive bursts is 25.6ns. The synchronisation pattern, burst delimiter, and end burst delimiter structures have been adopted from the IEEE Standard 802.3av, and they are all considered as part of the payload in measuring the burst BER. BERs were measured in a given packet, always preceded by the worst-case packet that maximally stressed the packet under consideration. For soft packets, the worst-case preceding packet is the loud packet. Similarly, for the loud packet, the worst-case preceding packet is the soft packet. The measured BERs are shown in Fig. 4b for the best-case and worst-case polarisations. For best-case polarisation, over 19dB dynamic range (assuming a BER less than 1.1x10−3) corresponding to the 10GEPON power levels is easily achievable. For practical reasons the maximum packet power level was limited to −6dBm for the measurements presented here. However, the trend of the best polarisation curve suggests that this dynamic range could be further improved, since for the loud packet there is still a significant margin before the FEC threshold is reached. A reduction to 14dB dynamic range in the worst-case polarisation can also be seen, which is attributed to the relatively large total PDG of the three SOAs used in the system. This can be improved in principle by using lower PDG SOAs in the reach-extender together with an APD receiver at the OLT (which avoids PDG of the third SOA).3. Conclusion
We have experimentally demonstrated a reach-extender for the upstream direction of 10Gb/s PONs. Up to 12dB compression of a 19dB input dynamic range was achieved. A reduction in patterning induced penalties for soft packets can be achieved by using the broadband ASE from the 1st stage SOA to clamp the gain of the 2nd stage SOA. For loud packets, it is shown that significant reduction in patterning induced penalties can be achieved through careful optimisation of the bias current of the 2nd stage SOA. The reach-extender is shown to be able to support 70km reach for a 32-split 10Gb/s PON.
Acknowledgments
This work was funded by Science Foundation Ireland under Grant 06/IN/I969
References and links
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